https://community.element14.com/products/manufacturers/wuerth-elektronik/The official Würth Elektronik element14 page with information, blogs and discussions within the electronics scene.en-USTelligent Community 12https://community.element14.com/products/manufacturers/wuerth-elektronik/f/forum/56565/red-led-occasionally-not-working-on-1312020030000-chip-led/232839Mon, 05 Jan 2026 17:42:00 GMT93d5dcb4-84c2-446f-b2cb-99731719e767:4416fe83-d581-4587-bf7a-d8b151a7d3ddmichaelkellettAs Shabaz has reminded you - 3.0V is out of spec for this device - there is no reason to expect it to work correctly. It will of course obey the usual laws of physics, but unless, you have detailed information about its insides, you can't predict how these will manifest themselves - it may make prefect sense that the red LED fails to illuminate when the chip temperature is just at a certain point. I don't understand which clock you are referring to - DIN on the diagram on the data sheet is a combined clock and data signal and the bit period should be between 0.9 and 1.5us - which is equivalent to 1.1MHz > fDIN > 0.667Mhz. Check your waveforms on a scope (and post them here if you are still stuck). MKhttps://community.element14.com/products/manufacturers/wuerth-elektronik/f/forum/56565/red-led-occasionally-not-working-on-1312020030000-chip-led/232838Mon, 05 Jan 2026 17:14:00 GMT93d5dcb4-84c2-446f-b2cb-99731719e767:02bfdd24-abe9-479c-beda-eaf3020a7ca2shabazHave you checked the datasheet, to see what the max data rate is supposed to be? Was 4 MHz chosen regardless? Unless it's a non-critical product (such as a toy or game) where there's no issue with a faulty indication, I think it's best to use the product within the specification, which means 3.3V minimum, not 3V.https://community.element14.com/products/manufacturers/wuerth-elektronik/f/forum/56565/red-led-occasionally-not-working-on-1312020030000-chip-ledMon, 05 Jan 2026 17:04:00 GMT93d5dcb4-84c2-446f-b2cb-99731719e767:7f850a99-64d2-4197-8014-b119aa6fd1d2DarienRodriguesWe have the WL-ICLED 1312020030000 chip LED powered from a 3V source. The red, green and blue LEDs independently work most of the time. But occasionally, the red LED stops working, while the blue and green continue to work. We are controlling the DIN signal from a MCU, which is also powered from the 3V DC source. I understand that the rated VDD supply is 3.3-5.5V. It does not make sense that only the red LED should stop working when the voltage is below the rated minimum. Additionally, this failure is only occasionally observed. Has anyone else faced this issue. If yes, how was this resolved? Should the DOUT pin be pulled low or connected to GND, if not used? We are using a 4 MHz clock frequency for the DIN signal.wl-icledchip LEDhttps://community.element14.com/products/manufacturers/wuerth-elektronik/b/blog/posts/isolated-rs-485-interface-based-on-4-channel-digital-isolator-with-integrated-dc-dc-converterMon, 01 Dec 2025 12:49:00 GMT93d5dcb4-84c2-446f-b2cb-99731719e767:b5257823-0369-4510-8609-31825049c2caWürth ElektronikSupport Note Isolated RS-485 interface based on 4-channel digital isolator with integrated DC/DC converter SN026 by Artem Beliakov This design example shows an isolated RS-485 interface using a 4-channel reinforced digital isolator with an integrated DC/DC converter, reducing component count and board space. The design is optimized for half-duplex operation at data rates up to 10 Mbps and meets CISPR 32 Class B radiated emission standard. 1. OVERVIEW OF RS-485 The RS-485 interface is widely used in industrial and commercial applications due to its robustness, long-distance data transmission capabilities, and resilience to electrical noise. RS-485 supports multiple devices on a single bus, making it ideal for complex systems and distributed networks. The main features of this interface include: Long-Distance Communication: Can transmit data up to 1200 meters at lower data rates (up to 100 kbps) Flexible Data Rates: Supports data rates from a few kbps up to 10 Mbps for short distances (~15 meters), with the data rate decreasing as the distance increases. However, with today's technology, signal rates beyond the specification of up to 50 Mbps are possible. Multi-Device Support: Allows up to 32 devices on a single bus (32-unit load (UL) or up to 256 low-unit-load transceivers where one transceiver is 1/8 UL), providing flexible connectivity for multi-node systems. Differential Signaling: Using differential signaling (balanced line), achieves high immunity to electrical noise, making it suitable for environments with significant EMI. Thanks to these benefits, RS-485 is widely used in industrial automation, building automation, energy management, motor control and renewable energy systems, where stable and noise-resistant data communication is essential for reliable operation in demanding environments. 2. IMPORTANCE OF RS-485 ISOLATION The differential nature of RS-485 signaling helps reduce noise, but it can still be affected by strong electromagnetic interference (EMI). For this reason, isolation is very important in modern applications where harsh conditions and high electrical noise levels are common. Isolated RS-485 provides extra protection and improves performance by separating the communication lines from the system ground, offering several benefits: Protection Against Ground Loops: Isolation prevents issues caused by ground potential differences, ensuring stable communication in large-scale systems with distributed equipment. Noise Immunity: Isolation (along with additional measures such as filtering, shielding and proper PCB layout) protects sensitive equipment from high levels of EMI and strong magnetic fields often found in industrial environments, helping maintain data integrity. Surge and Transient Protection: Galvanic isolation, complemented by surge protection devices (such as TVS diodes), allows to withstand voltage surges and transients, protecting components and improving the system's overall reliability. Safety in High-Voltage Applications: In systems such as energy meters and renewable energy, isolation helps protect low-voltage control systems from high-voltage sections, ensuring operator and equipment safety. Figure 1: Block diagram of isolated half-duplex RS-485 interface. 3. DESCRIPTION OF DESIGN EXAMPLE BOARD 3.1 Key parameters The board is optimized for a data rate of 10 Mbps and a shielded twisted cable length of 10 m (distance between the transmitter and receiver boards during radiated emission test). Table 1: Key parameters of isolated RS-485 board. The integrated DC/DC converter within the digital isolator cannot operate in boost mode therefore the output voltage must be less than or equal to the input voltage (VCC). In the design example the output voltage selection pin (SEL) of the digital isolator is connected to the ground (GND2) for 3.3 V output voltage. 3.2 Configuration The top view of the application example board is shown in Figure 2. The board can be conditionally divided into 8 blocks: Pins A and B: non-inverting and inverting bidirectional bus data lines (RS-485 input) Filter circuit of the data lines and TVS diodes for overvoltage protection Half-duplex RS-485 transceiver in SOIC-8NB package 4-channel digital isolator with integrated isolated DC/DC converter Filter circuit of the DC supply voltage line Terminal block screw connector for power supply THT 3-pin header for signals from controller SMA connectors for RS-485 interface signals* * The SMA connectors (high-speed connectors) are used for a test signal feed by a signal generator into the PCB. This is a simulation of a single ended board level signal source feeding the signal to the digital isolator. Figure 2: Top view of the application example board. Figure 3 shows the schematic of the transceiver and receiver boards used during the radiated emission test. The differences between the boards are detailed in Section 6.1. Figure 3: Schematic of isolated half-duplex RS-485 transceiver and receiver boards. 4. SELECTION OF COMPONENTS 4.1 Digital Isolator 18024115401H The 18024115401H is a 4-channel digital isolator with an integrated isolated DC/DC converter that provides up to 0.65 W of isolated power (for power supply of internal structure of the digital isolator and the half-duplex RS-485 transceiver) in a SOIC-16WB package. The availability of an integrated DC/DC converter significantly reduces the number of components in the design saving board space. In addition, the converter has integrated protection systems that guard against thermal overstress with thermal shutdown and protect against electrical damage using overcurrent, shortcircuit and undervoltage circuitry. The 18024115401H has a 3/1 channel configuration (3 forward and 1 reverse channels). This channel configuration is required for normal operation of the isolated RS-485 interface in half-duplex transmission. Two forward channels of the 18024115401H are used to isolate control signals from a microcontroller that switch the half-duplex RS-485 transceiver (RE and DE pins) between transmit and receive modes. The other two channels of the digital isolator (1 forward and 1 reverse) are used to isolate bus interface signals passing through the half-duplex RS-485 transceiver (DI and RO pins) in transmit and receive modes. Key Features of the 18024115401H: 4-channel digital isolator with integrated 0.65 W Isolated DC/DC converter; UL1577 recognized: 5000 V RMS isolation voltage for 60 s; DIN EN IEC 60747-17 (VDE 0884-17):2021-10 certified: Reinforced isolation; Maximum repetitive peak isolation voltage: 1414 V PK ; Maximum working isolation voltage: 1000 V RMS and 1414 V DC ; Maximum transient isolation voltage: 7070 V PK ; Maximum surge isolation voltage: 7070 V PK ; Input voltage range: 3.15 V to 5.5 V; User-selectable output voltage: 3.3 V or 5 V; Data rate up to 100 Mbps; ±150 kV/µs typ. CMTI; Complies with EN55032 (CISPR-32) class B conducted and radiated emissions standard (with the reference layout specified in the data sheet); Ambient temperature range: -40°C to 125°C. 4.2 Half-Duplex RS-485 Transceiver Half-duplex RS-485 transceivers provide bidirectional communication over a single pair of wires by alternating between transmitting and receiving data. This allows multiple devices to share the same communication line efficiently, reducing the number of required wires and simplifying system design. The design example uses a standard half-duplex RS485 transceiver in the SOIC-8NB package. Figure 4: Simplified block diagram of non-isolated half-duplex RS-485 transceiver in SOIC-8NB package. 4.3 Transient Protection The WE-TVS diode 824022 (two bidirectional TVS diodes in SOT-23 package) with a channel operating voltage of 5 V were selected to protect data line from overvoltage. The 824022 TVS diode has a very low input capacitance (15 pF), which helps minimize signal distortion, maintain high data rates and preserve the integrity of differential signals in highspeed RS-485 communication systems. 4.4 Filter circuit of the RS-485 data lines The WE-SL2 744222 common mode line filter was selected to the filter the RS-485 data lines. A detailed description of how to select the right components for the filter can be found in the Application Note ANP083 ‘’ Adapter PCB for filtering electromagnetic interference on an RS-485 interface ’’ (Section 2.1 Filter circuit of the data lines). 4.5 Filter circuit of the supply voltage line The WE-PD2 SMT power inductor 744773047 (4.7 µH) and WCAP-CSGP MLCC chip ceramic capacitors 885012209014 (10 µF/16 V, X7R, 1210) were selected for the input and output filter of the integrated DC/DC converter. For detailed information on filter selection and layout of the converter, refer to the 18024x15401x digital isolator datasheet (Section 20 Design example) and the online tool, REDEXPERT EMI Filter Designer . 5. REDUCING COMMON-MODE INTERFERENCE WITH OVERLAPPING STITCHING CAPACITANCE The 18024115401H digital isolator provides galvanic isolation between the input and output of the system, but parasitic coupling capacitance of an isolation barrier allows common mode currents to flow. As a result, the isolator may be a source of common-mode interference. Typically, design engineers address this issue using an external Y-capacitor between the input and output. However, an alternative approach is to utilize the parasitic capacitance between PCB layers as an integrated Y-capacitor. This method, commonly referred to as stitching capacitance, provides an effective high-frequency return path for common-mode noise, reducing EMI without requiring external components. Careful attention needs to be taken regarding the layout to meet any safety isolation standards. Depending on the targeted safety standard the requirements applying to inner layers for thickness and distance along a cemented joint. The outer layers and any exposed inner layer edges are subject to the creepage and clearance rules. This design example is designed to meet reinforced isolation. This design example board has a 4-layer PCB structure. The stitching capacitance is formed by the parasitic properties of overlapping copper planes in different PCB layers (between internal layer 1, internal layer 2 and bottom layer). The PCB layers and the overlap that form the stitching capacitance of the board are shown in Figure 5. Figure 5: Overlap area of isolated RS-485 design example board Figure 6: Layer stack legend. The parasitic capacitance between two overlapping PCB planes can be approximated by the parallel plate capacitance formula: (1) Where: C is the stitching capacitance in farads (F); ε 0 is the vacuum permittivity ≈ 8.854×10 −12 F/m; ε r is the relative dielectric constant of the PCB material (this information is provided by a PCB manufacturer, typically 4-5); A is the overlapping area of the planes in square meters (m²); d is the distance between the planes (dielectric thickness) in meters (m). Key parameters of the board for the stitching capacitance calculation: Overlap area (between GND1 and GND2 layers): 14 x 44 mm 2 = 616 mm 2 ; Distance between overlap layers and relative dielectric constant: Internal layer 1 to internal layer 2: 1.2 mm, ε r = 4.6; Internal layer 2 to bottom layer: 0.14 mm, ε r = 4. The total stitching capacitance of the PCB is the sum of the capacitances between the two overlapping regions: (2) The approximate stitching capacitance of the design is 177 pF, providing an effective high-frequency return path for common-mode noise and significantly improving EMI performance. Measurements of radiated emissions demonstrate the success of this approach (Figure 9), as the 4-layer board with stitching capacitance remains well below the required limits, proving that this integrated design effectively suppresses common-mode interference. More detailed information about common mode interference and coupling capacitance can be found in Application Note ANS022 . 6. TESTING 6.1 Test Setup The test setup for the radiated emission measurement is shown in Figure 7. The configuration differences between the transmitter and the receiver boards are: The transmitter board: resistors R4 and R5 are not installed. The receiver board: resistors R4 and R5 are installed. 6.2 Radiated Emission Measurements made with 10 m shielded twisted cable length between transmitter and receiver boards and 10 Mbps data rate. Figure 8: Transmitter board and signal generator board in EMC test chamber. Figure 9: Radiated emission (CISPR 32 Class B). 6.3 Propagation Delay The propagation delay between the input and output signals of the digital isolator is ~15 ns (Figure 10), ensuring fast switching, minimal signal distortion and stable communication via the RS-485 interface. Figure 10: Propagation delay between input (Channel 1) and output (Channel 2) signals of the digital isolator 18024115401H. A APPENDIX A.1 Bill of Material Designator Description WE series Order Code Manufacturer Quantity C1, C2 Filter ceramic chip capacitor 10 µF, 16 V, X7R, 1210 WCAP-CSGP 885012209014 Würth Elektronik 2 C3, C4 Ceramic chip capacitor 10 µF, 16 V, X7R, 1210 WCAP-CSGP 885012209014 Würth Elektronik 2 C5 Ceramic chip capacitor 100 nF, 100 V, X7R, 0805 WCAP-CSGP 885012207128 Würth Elektronik 1 C6, C7 Ceramic chip capacitor 10 pF, 25 V, NP0, 0603 WCAP-CSGP 885012006032 Würth Elektronik 2 R1 SMD resistor 50 Ω, 0.1 W, 0603 1 R2, R6 SMD resistor 10 Ω , 0.1 W, 0603 WRIS-RSKS 560112116013 Würth Elektronik 2 R3 SMD resistor 120 Ω , 0.1 W, 0603 WRIS-RSKS 560112116119 Würth Elektronik 1 R4,R5 SMD resistor 0 Ω , 0.1 W, 0603 WRIS-RSKS 560112116001 Würth Elektronik 2 U1 4-channel digital isolator with integrated DC/DC, SOIC-16WB WPME-CDIP 18024115401H Würth Elektronik 1 U2 Half-duplex RS-485 transceiver 50 Mbps, SOIC-8NB 1 D1 2-channel TVS Diode, 5 V, 12 pF, SOT23-3L WE-TVS 824022 Würth Elektronik 1 L1 Filter SMD inductor 4.7 µH, 4532 WE-PD2 744773047 Würth Elektronik 1 L2 SMT common mode line filter 1000 µH, 0.8 A, 80 V WE-SL2 744222 Würth Elektronik 1 J1 THT horizontal entry modular, pitch 5 mm, 2p WR-TBL 691502710002 Würth Elektronik 1 J2, J3 SMA PCB end launch connector WR-SMA 60312202114509 Würth Elektronik 2 P1 THT 3-pin header, vertical, single row, pitch 2.54 mm WR-PHD 61300311121 Würth Elektronik 1 P2 THT 1-pin header, vertical, single row, pitch 2.54 mm WR-PHD 61300111121 Würth Elektronik 1 A.2 Supporting design file archive Supporting design file archive contains Support Note, Schematic, Bill of materials, Gerber files, NC Drill files, Support Note, Layer Definition and Layer Stack Legend. The link to the file archive is available: https://www.we-online.com/components/products/media/860161 IMPORTANT NOTICE The Application Note is based on our knowledge and experience of typical requirements concerning these areas. It serves as general guidance and should not be construed as a commitment for the suitability for customer applications by Würth Elektronik eiSos GmbH & Co. KG. The information in the Application Note is subject to change without notice. This document and parts thereof must not be reproduced or copied without written permission, and contents thereof must not be imparted to a third party nor be used for any unauthorized purpose. Würth Elektronik eiSos GmbH & Co. KG and its subsidiaries and affiliates (WE) are not liable for application assistance of any kind. Customers may use WE’s assistance and product recommendations for their applications and design. The responsibility for the applicability and use of WE Products in a particular customer design is always solely within the authority of the customer. Due to this fact it is up to the customer to evaluate and investigate, where appropriate, and decide whether the device with the specific product characteristics described in the product specification is valid and suitable for the respective customer application or not. The technical specifications are stated in the current data sheet of the products. Therefore the customers shall use the data sheets and are cautioned to verify that data sheets are current. The current data sheets can be downloaded at www.we-online.com. Customers shall strictly observe any product-specific notes, cautions and warnings. WE reserves the right to make corrections, modifications, enhancements, improvements, and other changes to its products and services. WE DOES NOT WARRANT OR REPRESENT THAT ANY LICENSE, EITHER EXPRESS OR IMPLIED, IS GRANTED UNDER ANY PATENT RIGHT, COPYRIGHT, MASK WORK RIGHT, OR OTHER INTELLECTUAL PROPERTY RIGHT RELATING TO ANY COMBINATION, MACHINE, OR PROCESS IN WHICH WE PRODUCTS OR SERVICES ARE USED. INFORMATION PUBLISHED BY WE REGARDING THIRD-PARTY PRODUCTS OR SERVICES DOES NOT CONSTITUTE A LICENSE FROM WE TO USE SUCH PRODUCTS OR SERVICES OR A WARRANTY OR ENDORSEMENT THEREOF. WE products are not authorized for use in safety-critical applications, or where a failure of the product is reasonably expected to cause severe personal injury or death. Moreover, WE products are neither designed nor intended for use in areas such as military, aerospace, aviation, nuclear control, submarine, transportation (automotive control, train control, ship control), transportation signal, disaster prevention, medical, public information network etc. Customers shall inform WE about the intent of such usage before design-in stage. In certain customer applications requiring a very high level of safety and in which the malfunction or failure of an electronic component could endanger human life or health, customers must ensure that they have all necessary expertise in the safety and regulatory ramifications of their applications. Customers acknowledge and agree that they are solely responsible for all legal, regulatory and safety-related requirements concerning their products and any use of WE products in such safety-critical applications, notwithstanding any applicationsrelated information or support that may be provided by WE. CUSTOMERS SHALL INDEMNIFY WE AGAINST ANY DAMAGES ARISING OUT OF THE USE OF WE PRODUCTS IN SUCH SAFETYCRITICAL APPLICATION. DIRECT LINK SN026 | Isolated RS-485 interface based on 4-channel digital isolator with integrated DC/DC converter USEFUL LINKS: Application Notes : https://we-online.com/en/support/knowledge/application-notes Services: https://we-online.com/en/products/components/service Contact : https://we-online.com/en/support/contact CONTACT INFORMATION Würth Elektronik eiSos GmbH & Co. KG Max-Eyth-Str. 1, 74638 Waldenburg, Germany Tel.: +49 (0) 7942 / 945 – 0 Email: appnotes@we-online.de Web: https://www.we-online.comwurth_electronicssupport notewürth elektronikknowledgewurth_elektronikhttps://community.element14.com/products/manufacturers/wuerth-elektronik/b/blog/posts/taking-a-closer-look-at-mechanical-components-and-cleaning-processesThu, 27 Nov 2025 11:56:00 GMT93d5dcb4-84c2-446f-b2cb-99731719e767:779d9041-fd0a-40bd-b7a6-24d386329d80Würth ElektronikTaking A Closer Look at Mechanical Components and Cleaning Processes By Jasmin Hsu Over the years, we have frequently received questions regarding the washing process of mechanical switches. We have observed that washing has become a standard part of the production process for customers. Washing the PCB helps remove residues on the PCB to avoid corrosion, short circuit or maybe the customer has afterward another coating or potting processes. And in the market, we also see different definitions like dustproof, waterproof, IP-rated, washable. But in most of the times, manufactures still do not allow customer to wash their mechanical components. The second characteristic numeral indicates the degree of protection provided by equipment against ingress of water. The tests for the second characteristic numeral, except for numeral 9, are carried out with fresh water at temperature range of 15 °C to 35 °C. The actual protection may not be sufficient if cleaning operations involve high water temperatures outside the requirements of the second characteristic numeral and/or cleaning solvents are used. So far, based on what we know, there isn't a single standard for the cleanability of general electronic components. One of the reasons is that these standards (AEC standards, IEC standards) ensure component reliability under various conditions, but cleanability is not their primary focus. IPC standards define cleanliness requirements for printed circuit boards (PCBs) and assembled boards, but these standards apply for cleaning after manufacturing, not for the component's ability to withstand cleaning. Why might mechanical parts be not suitable for the washing process? We need to break this question into 2 aspects. When you search for IPx7, you can find a lot of information related to IEC 60529. Behind each characteristic numeral, there is a definition of how the enclosure must be tested with water (e.g. spray, submersion). It is important to note is that: the test is performed using only water. However, in the actual production processes, the cleaning/washing process is carried out using a combination of cleaning agent, defoamer & water. Different water temperature and water pressures depending on actual practice. So, if the test conditions of IEC 60529 are so different from the actual cleaning process, can we really say that IPx7 parts are Washable? From another perspective, the term “mechanical switches” is already quite self-explanatory.. They are built from various mechanical spare parts to become a mechanical switch. As an example, imagine stacking Lego bricks on each other. If you put them into water, you will see a lot of water bubbles will be popping up indicating there are air gap between the Lego bricks. Or if you build a house without putting cement between the bricks, wind and water will pass easily into the house, right? With the two examples above, we hope we can give you a deeper dive into our today’s topic. To build a house, we need bricks and cement to seal the air gaps. Similarly, for a mechanical switch, we need also some additional protection or ingredients to the component to make it more protected from fluid ingress. And recent solution is to use silicone rubber no matter in which form or shape and the function of the rubber is to is to be designed in such a way that it closes or minimizes the air gap between the spare parts But does using silicone rubber automatically mean the switch is airtight ? Take a look on these two pictures: they show two different internal structures both using silicone rubber inside a switch. One silicone rubber is placed on the gap between the cover and the base., In the other , it is placed above the stationary contact to protect it . Which one do you think performs better when water flows from top to bottom? (Picture on right:, air gap on the red line areas between cover and base) So, even when silicone rubber is used, it does not mean the switch is sealed. What we want to express is that, that even with silicone rubber inside the switch, without a proper design to close or tighten the possible air gap it is not possible to achieve the sealing function for a mechanical switch. Conclusion : IP testing is performed using only water, which is different from what is used in the actual cleaning process. Usage of silicone rubber improves protection but does not mean it is sealed.switcheswurth_electronicswürth elektronikknowledgewurth_elektronikhttps://community.element14.com/products/manufacturers/wuerth-elektronik/w/documents/27409/ane004-redfit-idc-skedd-connectorWed, 26 Nov 2025 11:23:00 GMT93d5dcb4-84c2-446f-b2cb-99731719e767:a4df3b72-b4c3-45b7-b81e-23659590abb0Würth ElektronikAPPLICATION NOTE Please find a updated new Application Note here: ANE017 | REDFIT SKEDD Crimp Connector USEFUL LINKS ANE017 | REDFIT SKEDD Crimp Connector Application Notes www.we-online.com/appnotes REDEXPERT Design Plattform www.we-online.com/redexpert Toolbox www.we-online.com/toolbox Produkt Catalog www.we-online.com/products CONTACT INFORMATION appnotes@we-online.com Tel. +49 7942 945 - 0 Würth Elektronik eiSos GmbH & Co. KG Max-Eyth-Str. 1 ⋅ 74638 Waldenburg ⋅ Germany www.we-online.comconnectorsfirmwaredebugredfitconnector_applicationApplication Noteswurth_elektronikhttps://community.element14.com/products/manufacturers/wuerth-elektronik/w/documentsWed, 26 Nov 2025 11:23:00 GMT93d5dcb4-84c2-446f-b2cb-99731719e767:4cfe381d-1b58-4ac1-972c-e6fdee5bf897https://community.element14.com/products/manufacturers/wuerth-elektronik/b/blog/posts/redfit-skedd-crimp-connector-plugthepowerMon, 24 Nov 2025 12:08:00 GMT93d5dcb4-84c2-446f-b2cb-99731719e767:043fc13f-43ee-4de6-a43b-bae531396ca7Würth ElektronikApplication Note REDFIT SKEDD Crimp Connector #PLUGTHEPOWER ANE017 by Moritz Jakob & Andreas Aigner 01. INTRODUCTION SKEDD is a technology developed by Würth Elektronik that allows connectors to contact directly with the printed circuit board without soldering. The combination with the crimp technology of the REDFIT Crimp SKEDD Connector offers significant advantages over soldered connectors. 02. CORE OF THE SKEDD TECHNOLOGY The core of SKEDD technology consists of innovatively designed stamped contacts that provide an electrically stable connection directly within the plated through-holes of the printed circuit board. The mechanical preload generated during contact insertion ensures a stable electrical connection even under high vibration conditions. The use of high-performance alloys as contact materials enables a high normal contact force, which ensures reliable electrical connectivity under demanding mechanical conditions. Thus, SKEDD technology provides a stable electrical connection. Difference from Press-Fit Technology with Solid Pins In conventional press-fit connections using solid pins, the high forces applied during the joining process create a permanent, non-detachable connection between the contact partners. The plated through-hole of the printed circuit board adapts to the contact because of the press-in process. With SKEDD technology, no permanent deformation of the plated through-hole occurs. Only the SKEDD forks are preloaded and return to their original position after the connection is released. In other words, SKEDD utilizes the elastic range of the copper material (see Figure 1). A sufficiently large elastic range is essential for compensating hole and positional tolerances while maintaining an adequate normal contact force. This range is influenced by the selection of suitable materials and by a design that provides an optimal balance between stiffness and elasticity. Figure 1: Tension and elongation behavior of copper material 03. SKEDD AS A CRIMP CONTACT With the REDFIT Crimp SKEDD Connector, this technology has been further developed for use in PCB power supply applications to achieve higher current-carrying capacity while maintaining high contact stability. In addition to improved electrical properties, the mechanical stability has also been raised to a new level. Thanks to SKEDD technology, both inserting the contact into the housing and the connector into the printed circuit board can be done manually without tools. This eliminates the need for both the soldering process and a mating connector. For the connection to the wire, well-established crimping technology is used. Figure 2 shows the REDFIT Crimp system with contact and connector in the unmated condition. Figure 2: SKEDD Crimp contact 04. ELECTRICAL PARAMETERS As mentioned previously, the new REDFIT version is designed for power supply applications on the printed circuit board. With a current-carrying capacity of 16 A at an operating voltage of 400 VAC, it is also ideally suited for mains power applications. Figure 3: REDFIT Crimp for power supply 05. MECHANICAL STABILITY In industrial applications, mechanical stability is of high importance in addition to electrical properties. Thanks to the newly developed and patented mechanical locking mechanism, forces up to 10 g can be withstood without mechanical or electrical failure. Conventional connectors are typically tested at 5 g over frequencies from 10 to 55 Hz. Figure 4 shows the process of how the connector is tested for vibration. The acceleration acts on the connector for a duration of 7.5 hours with up to 10 g. This ensures that no mechanical or electrical failures occur under high mechanical stress. A sinusoidal frequency sweep from 10 to 2000 Hz is tested. Figure 4: Vibration curve 10 g (10 to 2000 Hz) During the development of the active locking mechanism, special emphasis was placed on user friendliness. After manually inserting the REDFIT connector into the printed circuit board, the collar is simply pushed down to its lowest position. In this position, a locking dome secures the latching tabs in the PCB, providing both tactile and audible feedback to the user. This ensures stable mounting, making the connector suitable even for mechanically demanding applications. Figure 5: Positive feedback when locking with slider The patented locking mechanism of the connector on the printed circuit board combines the mechanical and electrical connection within a single plated through-hole. This saves space on the PCB, allowing for a more compact connector design. As a result, the REDFIT Crimp also meets the requirements of miniaturization. Figure 6: Locked connector An additional plastic pin prevents accidental reverse polarity of the contacts. An extra marking on the housing indicates the position of PIN 1 in ascending order. Figure 7: Plastic pin assures correct polarity. 06. PCB LAYOUT The REDFIT Crimp connector requires a PCB thickness of 1.6 mm. For optimal electrical and mechanical connection, the tolerances used for press-fit technology should be applied during PCB manufacturing. The PCB plating can be done with chemical tin, ENIG, or HAL. The pad size is negligible, as the contact is formed within the sleeve. However, the pad is required for PCB production. Thus, the annular ring size can be minimized. The drill hole specifications are shown in Figure 8. The plated through-hole should have a final diameter of 3.5 mm. Figure 8: Drill hole and plating specification (in mm) Figure 9 shows the recommended layout for a REDFIT Crimp with three pins. Since the mechanical fixation also occurs within the plated through-hole, no additional mounting points are required. Depending on the application, the REDFIT Crimp can also be mounted in the opposite insertion direction. In this case, the layout simply needs to be placed on the desired side of the PCB. Figure 9: PCB Layout (in mm) 07. AVAILABLE VERSIONS The REDFIT Crimp is available in configurations from 1 to 5 pins. Visual coding can be achieved using differently colored connector collars. The collars are available in red, blue, yellow, black, and white. Figure 10: Color-Coded to the number of pins 08. CONCLUSION The REDFIT Crimp SKEDD connector can be used flexibly in mechanically and electrically demanding applications. It is mounted directly onto the printed circuit board by hand, eliminating the need for both the soldering process and a mating connector. SKEDD technology can supply current of up to 16 A to fully SMT-populated PCBs. Depending on the layout, mounting in the opposite insertion direction is also possible. The entire SKEDD product range at a glance. I M P O R T A N T N O T I C E The Application Note is based on our knowledge and experience of typical requirements concerning these areas. It serves as general guidance and should not be construed as a commitment for the suitability for customer applications by Würth Elektronik eiSos GmbH & Co. KG. The information in the Application Note is subject to change without notice. This document and parts thereof must not be reproduced or copied without written permission, and contents thereof must not be imparted to a third party nor be used for any unauthorized purpose. Würth Elektronik eiSos GmbH & Co. KG and its subsidiaries and affiliates (WE) are not liable for application assistance of any kind. Customers may use WE’s assistance and product recommendations for their applications and design. The responsibility for the applicability and use of WE Products in a particular customer design is always solely within the authority of the customer. Due to this fact it is up to the customer to evaluate and investigate, where appropriate, and decide whether the device with the specific product characteristics described in the product specification is valid and suitable for the respective customer application or not. The technical specifications are stated in the current data sheet of the products. Therefore the customers shall use the data sheets and are cautioned to verify that data sheets are current. The current data sheets can be downloaded at www.we-online.com. Customers shall strictly observe any product-specific notes, cautions and warnings. WE reserves the right to make corrections, modifications, enhancements, improvements, and other changes to its products and services. WE DOES NOT WARRANT OR REPRESENT THAT ANY LICENSE, EITHER EXPRESS OR IMPLIED, IS GRANTED UNDER ANY PATENT RIGHT, COPYRIGHT, MASK WORK RIGHT, OR OTHER INTELLECTUAL PROPERTY RIGHT RELATING TO ANY COMBINATION, MACHINE, OR PROCESS IN WHICH WE PRODUCTS OR SERVICES ARE USED. INFORMATION PUBLISHED BY WE REGARDING THIRD-PARTY PRODUCTS OR SERVICES DOES NOT CONSTITUTE A LICENSE FROM WE TO USE SUCH PRODUCTS OR SERVICES OR A WARRANTY OR ENDORSEMENT THEREOF. WE products are not authorized for use in safety-critical applications, or where a failure of the product is reasonably expected to cause severe personal injury or death. Moreover, WE products are neither designed nor intended for use in areas such as military, aerospace, aviation, nuclear control, submarine, transportation (automotive control, train control, ship control), transportation signal, disaster prevention, medical, public information network etc. Customers shall inform WE about the intent of such usage before design-in stage. In certain customer applications requiring a very high level of safety and in which the malfunction or failure of an electronic component could endanger human life or health, customers must ensure that they have all necessary expertise in the safety and regulatory ramifications of their applications. Customers acknowledge and agree that they are solely responsible for all legal, regulatory and safety-related requirements concerning their products and any use of WE products in such safety-critical applications, notwithstanding any applications-related information or support that may be provided by WE. CUSTOMERS SHALL INDEMNIFY WE AGAINST ANY DAMAGES ARISING OUT OF THE USE OF WE PRODUCTS IN SUCH SAFETYCRITICAL APPLICATIONS. DIRECT LINK ANE017 | REDFIT SKEDD Crimp Connector USEFUL LINKS: Application Notes : https://we-online.com/en/support/knowledge/application-notes Services: https://we-online.com/en/products/components/service Contact : https://we-online.com/en/support/contact CONTACT INFORMATION Würth Elektronik eiSos GmbH & Co. KG Max-Eyth-Str. 1, 74638 Waldenburg, Germany Tel.: +49 (0) 7942 / 945 – 0 Email: appnotes@we-online.de Web: https://www.we-online.comwurth_electronicswürth elektronikpowerapplication_noteskedd connectorwurth_elektronikhttps://community.element14.com/products/manufacturers/wuerth-elektronik/b/blog/posts/sn024-transient-suppression-at-different-interfacesTue, 18 Nov 2025 12:07:00 GMT93d5dcb4-84c2-446f-b2cb-99731719e767:0114dc76-a199-4fea-9820-84969f1cf8c9Würth ElektronikApplication Note Transient suppression at different interfaces SN024 by Nadine Simpfendoerfer, Swarnasree Banik 1. INTRODUCTION Due to the increasing number of electronic devices on the public power grid and network communication via data lines, measures to prevent mutual interference caused by electromagnetic interference are becoming increasingly important. Protecting the wired interfaces against transient overvoltage is particularly important. Filter circuits, which are placed at the interfaces of the devices, not only attenuate high-frequency emissions, and thus maintain radio protection, but reduce transient overvoltage directly at the input of a device so that the functions of the devices are not impaired. Basically, the following interface variants can be distinguished, each of which requires similar filter topologies: Grid interfaces, i.e. 230 V mains connection. Asymmetrical signal interfaces. Symmetrical signal interfaces. In this context, grid interfaces mean the interfaces between the power grid and consumer’s equipment. Here, line filters are used that attenuate interference emissions from devices into the power grid and protect a device from interference from the grid. Signal interface filters are designed for lower operating currents and voltages. The requirements for the filters differ in the type of signal transmission, which can be symmetrical (balanced) over two data lines or asymmetrical (unbalanced) via one data line and ground. Although symmetrical interfaces are characterized by lower interference emissions and higher interference immunity, the signal quality is affected by transient interference signals, differential-mode interference and common-mode interference. This support note describes the structure of various interface filters that help ensure signal integrity and protect against transients. Circuit diagrams of the filter circuits, information on component selection and practical examples enable easy transfer to other network or signal interfaces. The asymmetrical signal interface is described first in the following section. 2. EMI FILTER OF AN ASYMMETRICAL SIGNAL INTERFACE WITH TRANSIENT PROTECTION While unbalanced signal interfaces only use one data line per signal, interference emissions and interference immunity requirements are generally higher than with balanced signal interfaces. Figure 1 shows a possible filter circuit for attenuating interference emissions and ensuring interference immunity. Figure 1: Unbalanced signal interface filter. A digital (square wave) signal with amplitude of 5 V PP and frequency of 5 MHz on the data line is considered below. The standard value of 90 Ω is assumed for the impedance of long data signal lines. There is a voltage of 5 V DC at 2 A on the supply voltage line. Since the requirements for the filter circuits for the data line and the supply voltage line differ, the filtering of the data line will be described first. 2.1 Filter circuit of the data line To attenuate high-frequency interference on the data line, the two impedances Z 1 and Z 2b in Figure 2 form a low-pass π filter with the inductor. Protection against transient overvoltage is ensured by a transient protection device Z 2a . Figure 2: Filter on the data line of an unbalanced signal interface. The component selection is explained below using the example of the signal driver described above. Step 1: Determination of L 1 First, L1 is dimensioned and selected, where an SMT ferrite or an RF–ferrite can be used. Further information on the use of inductors or SMT ferrites is given in ANP025 and in ANP129 . The ohmic, or resistive, component of the impedance of the SMT ferrite should be as low as possible up to the fifth harmonic of the signal frequency, and the impedance at this frequency should be less than one-tenth of the system impedance to avoid affecting the useful signal. In our example, the fifth harmonic is at 25 MHz, as the signal frequency is a 5 MHz square wave. Since the system impedance in the example is 90 Ω, the impedance at 25 MHz should be at most about one-tenth, i.e., no more than 9 Ω. Figure 3 shows a typical impedance curve as seen in the data sheets. The proper inductors, in the example a SMT-ferrite can be selected with the help of REDEXPERT . Figure 3: Impedance characteristics WE-CBF SMT ferrite 74279273 . From Figure 3, it can be concluded that WE-CBF 74279273 SMT ferrite can be used. Step 2: Determination of Z 1 Z 1 is a capacitor that can be optionally used to attenuate interference frequencies in the RF range above 500 MHz originating from the driver. Based on practical experience, a desired attenuation of 10 dB should be achieved at 500 MHz, resulting in a voltage ratio of V OUT /V IN = 0.316. The capacitor impedance Z 1 (capacitive reactance) can be determined using the voltage divider in Figure 4. Figure 4: Voltage divider for calculating the capacitor Z 1 (1) Where: V OUT is the voltage across the capacitor Z 1 (V). V IN is the total input voltage to the circuit (V). V OUT /V IN is the voltage ratio from the attenuation. R i is the internal resistance of the signal source or system impedance (Ω). Z 1 is the capacitor impedance (capacitive reactance) (Ω). Given R i = 90 Ω, the calculated Z 1 from (1) is approximately 42 Ω. Next, the capacitance value (C Z1 ) required for attenuation at f = 500 MHz can be calculated as follows: (2) Substituting the known values in (2), Based on these calculations, the WCAP-CSGP 885012005040 MLCC capacitor with a value of 10 pF and a voltage withstand rating of 25 V is suitable. This capacitor is an NP0 ceramic type, offering more stable voltage and temperature behavior compared to X7R ceramics. When considering critical factors such as signal integrity due to the cutoff frequency of the filter and the capacitive load, it's important to account for parasitic capacitances of the conductor tracks and, if necessary, interface connectors. Step 3: Determination of Z 2 Z 2 consists of a component for transient protection (Z 2a ) and a capacitor (Z 2b ), whereby the parasitic capacitance of the transient protection device such as a TVS diode or a SMT varistor additionally contributes to the capacitance of the π filter. If the capacitance of the transient protection is high enough to achieve the required attenuation of 3 dB at 25 MHz (V OUT /V IN = 0.707), the capacitor may not be needed. The advantages of a varistor compared to a TVS diode are the higher parasitic capacitance, a shorter response time and the ability to absorb more energy with the same component size. The procedure for selecting Z 2 is the same, using the following voltage divider with V OUT /V IN and Z L1 : (3) Where: V OUT /V IN is the voltage ratio from the attenuation. Z L1 is the impedance of the ferrite at 25 MHz with Z L1 = 10 Ω. Z 2 represents the total impedance of Z 2a and Z 2b (Ω). From this, we calculate the impedance and capacitance required for the Z 2 : Impedance: Z 2 ≈ 25 Ω Therefore, capacitance needed: C Z2 ≈ 250 pF The operating voltage of the varistor must be greater than the maximum operating voltage of the system, which here is 5 V. Considering a safety buffer of 15%, the maximum operating voltage of the varistor is V DC > 5.75 V. Based on this value and the available space in the application, an SMT multilayer varistor is selected, which serves as the basis for the decision. In this case, the WE-VS 82536040 (Table 1) varistor is selected, whose operating voltage (V DC ) is 5.5 V and is therefore slightly below the required operating voltage. Once the varistor has been selected, the following calculations determine whether the correct selection has been made or not. According to the IEC/EN 61000-4-5 standard, the test pulse for surge between line and ground is V SURGE = 0.5 kV and the source impedance is 42 Ω. Table 1: Electrical properties of WE-VS 82536040 . To simplify our calculation, we specify the clamping voltage during the current surge as twice the breakdown voltage V BR . This value (2 x 8 = 16 V vs. clamping at 21 V) is slightly lower than the actual clamping voltage of the varistor at the surge current. Approximation: V CLAMP ~ 2 V BR . The varistor voltage (V VAR or V BR ) can be found in the varistor data sheet (see Table 1). The peak varistor current can be calculated as follows: (4) Where: I CLAMP, MAX is the peak current of the varistor (A). V SURGE is the test pulse voltage for surge (V). V BR is the varistor voltage or breakdown voltage at 1 mA (V). Z SOURCE is the source impedance as per IEC/EN 61000-4-5 (Ω). Substituting the known values in (4), Considering the varistor tolerance of 25%, the maximum clamping current is 11.62 A and is therefore below the maximum peak current of the varistor (30 A). Energy consumption and power loss calculation: To simplify our calculation, we assume that the pulse is a square signal with a pulse duration of 20 µs and the maximum clamping current of 11.6 A. The approximate energy can be calculated with twice the breakdown voltage and I CLAMP, MAX : (5) Where: I CLAMP, MAX is the maximum clamping current. V BR is the varistor voltage or breakdown voltage at 1 mA (V). W MAX is the energy consumption of the varistor (J). Substituting the known values in (5), For the example above, the SMT Varistor WE-VS 82536040 with a capacitance of 200 pF for Z 2a can be used. To achieve the necessary capacitance, an additional capacitor with 47 pF in size 0805 is also used for Z 2b in this case. The selected capacitor is the WCAP-CSGP 885012007080 . 2.2 Filter circuit of the supply voltage line In this section, the filter for the supply voltage cable entering the device is developed as shown in Figure 5. Figure 5: Power supply line filter of an unbalanced signal interface. Generally, transients are filtered at interfaces against the housing - shown with the “housing” symbol. To enable a connection between the negative wire (GND of the USB cable line) to the functional ground (GND of the circuit board) and still attenuate interference coming from the cable, an SMT ferrite is used between the housing ground and GND. The capacitor Z5b and the parasitic capacitance of Z5a drain RF emissions with respect to transient voltages coming from the cable to the housing ground. Step 1: Determination of Z 3 A decoupling capacitor with 100 nF is used as standard here. A X7R ceramic capacitor with sufficient high rated voltage should be used here, so as not to run into the voltage bias effect and to lose capacitance. Step 2: Determination of Z 4 and Z 5 Z 4a is a SMT varistor with C Ch = 200 pF. To achieve the necessary capacitance for the filter, an additional capacitor of 100 nF is usually used as standard for Z 4b , which should be the same capacitor as Z 3 .and Z 5b . Varistors Z 5a and Z 4a should be the same component. Step 3: Determination of L 2 and L 6 L 2 and L 6 are SMT ferrites, where the rated current should be considered first. This must be greater than 2 times the output current. In this example corresponds to 2 · 2 A = 4 A. Minimum rated current of SMT ferrite: I r = 4 A Supply voltage lines have no cutoff frequency and a very low source impedance of approx. Z OUT ≈ 2 ··· 9 Ω. As the noise source is the +V line, all other signal sources on the electronic board will create RF noise to the +V network which must be considered. This means that the impedance Z L2 or Z L6 should be as high as possible across the entire signal frequency range (typically 30 MHz to 1 GHz) to achieve sufficiently strong attenuation. The selection of the proper SMT ferrite should be done by REDEXPERT to get the right impedance over the relevant frequency range. The ferrite selected here is the WE-CBF SMT EMI Suppression Ferrite Bead 74279252 . The impedance chart is shown in Figure 6 and the electrical properties from the data sheet are shown in Table 2. Figure 6: Impedance characteristics of SMT ferrite CBF 74279252 Table 2: Electrical properties of the SMT ferrite CBF 74279252 . Remark to Electrical Grounding Unlike earthing for electrical safety in the sense of the Low Voltage Directive, the ground is a reference point for high-frequency signals in terms of EMC. Accordingly, the reference ground must have a low impedance and be free of high frequency "offset signals" and provide a reference point for filters and TVS components. The best reference is a very conductive metal housing. 3. EMI FILTER OF A SYMMETRICAL SIGNAL INTERFACE WITH TRANSIENT PROTECTION Balanced signal interfaces use differential signal pairs to transmit information and to attenuate common-mode interference. However, these interfaces are still susceptible to radiated electromagnetic interference and the coupling of transient voltages and currents. To reduce unwanted interference that can affect signal quality and system operation, additional filtering of the interface is essential. A good EMI filter reduces electromagnetic emissions from the signal source to the cable and vice versa, limits transient external interference to a tolerable level and has only a negligible effect on the signal integrity of the interface. Figure 7 depicts a possible filter circuit for a balanced signal interface. The circuit utilizes a soft low-pass filter to pass the signal in differential mode while attenuating common-mode components. Figure 7: Balanced signal interface filter. The symmetric low-pass filter allows differential-mode signals, which carry the essential information, to pass through unimpeded. Conversely, common-mode signals, which are noise components of the signal, are attenuated or damped by the filter. TVS-diode array provides transient voltage protection. These spikes are sudden, unwanted bursts of high voltage that can damage electronic components. When a transient overvoltage occurs, the TVS diodes limit the voltage and divert the transient impulse current from the circuit to ground (housing). The next section provides a practical example of component selection for a USB 2.0 interface. The USB 2.0 interface is a widely used for data and power transfer between electronic devices. It provides a signal bandwidth of approximately 240 MHz (480 Mbps) and an impedance of 90 Ω. A typical power supply for USB 2.0 is +5 V with a maximum current of 1 A. 3.1 Component Selection for USB 2.0 Interface Step 1: Determination of L 1 L 1 is an inductor with high impedance for common-mode noise that allows differential signals to pass with minimal attenuation in the relevant signal frequency range. L 1 selection is crucial in terms of differential and common-mode impedance needs. To ensure proper signal transmission, choke L 1 should be selected according to the interface type and datasheet information, taking the following parameters into account: System/Interface Impedance (Z SYSTEM ): Z SYSTEM can influence EMI filter design. Obtain the system or interface impedance from the data sheet or specifications. Differential-mode Impedance (Z DM ): The choke impedance in the differential mode should be lower than 1/10 of the system impedance (Z SYSTEM ) at the highest frequency within the signal bandwidth. This ensures minimal attenuation of the desired signals. Calculate the maximum allowed impedance for L 1 in the differential mode: (6) Common-mode Impedance (Z CM ): Z CM should be as high as possible, especially at frequencies corresponding to expected interference. At center frequency Z CM should be higher than Z SYSTEM . Select a choke (L 1 ) with a high Z CM , especially at frequencies of expected interference. Pay attention to the frequency response, current handling capability, and other relevant parameters of the choke. A suitable choke with a differential-mode impedance less than Z DM, MAX and a high common-mode impedance Z CM for our application is the WE-CNSW HF SMT common-mode line filter 744233900 . Step 2: Determination of Z 1 , Z 2 , Z 5 and Z 6 Capacitors (Z 1 , Z 2 ) and matching resistors (Z 5 , Z 6 ) form a "soft" low pass filter to limit signal harmonics or HF noise produced by the controller, resp. signal transceiver. The values of the resistors depend on the signal controller and its imbalance/mismatch and must be selected according to the measurement results. The values of Z5, Z6 are typically between 3 and 10 Ω. The capacitor values must be selected accordingly, the values vary in the range of 2.2 to 10 pF. Typical value for USB 2.0: 4.7 pF, e.g., WCAP-CSGP Ceramic Capacitor 885012005038 . The resistors Z 5 , Z 6 (e.g., 3.3 Ω) are typically used for reflection damping and filtering in conjunction with the capacitors (Z 1 , Z 2 ) to improve signal integrity and reduce interference. RF suitable types with a small body size should be used here. Step 3: Determination of Z 3 , Z 4 , L 2 , Z 7 and Z 8 TVS diodes (Z 3 , Z 4 ) and components like L 2 , Z 7 , and Z 8 provide transient protection against voltage spikes and surges. To properly handle transient currents, consider the minimum rated current (I r ) of components such as L 2 and the SMT ferrite (Z9a). L 2 , Z 7 , and Z 8 are used for low cut-off frequency filtering, typically less than 1 MHz. SMT Ferrite Bead (L 2 ): Used for low cut-off frequency. Ensure the choke has an appropriate minimum rated current Ir. To achieve broadband noise reduction, the SMD ferrite WE-CBF SMT EMI Suppression Ferrite Bead 742792032 (see Table 3). Table 3: Electrical properties of the SMT ferrite CBF 742792032 . Capacitors and Varistors (Z 7 , Z 8 ): Z 7 is a capacitor for low cut-off frequency filtering and Z 8 is a SMT varistor for transient protection. An example of a suitable capacitor for this application is WCAP-CSGP Ceramic Capacitor 885012206120 and for a varistor, WE-VS SMT Varistor 82536040 . TVS Diodes Array (Z 3 , Z 4 ): TVS diode arrays (Transient Voltage Suppression Arrays), such as the WE-TVS diode 8240136 from the "High Speed" series, are used to protect signal lines against transient voltages. It is important to note that these components also exhibit parasitic capacitance, which must not be too high relative to the data transfer rate or signal bandwidth. Information regarding suitable applications can be found in the datasheets. Step 4: Determination of Z 9a and Z 9b For high-frequency decoupling between housing and cable ground (GND), SMT ferrites (L 10 ) and SMT varistors ( Z 9a and Z 9b ), e.g., WE-CBF SMT EMI Suppression Ferrite Bead and WE-VS SMT varistor are used in combination with capacitors. This filter section helps reduce common-mode interference and improves system stability. Ensure that the current-carrying components are compatible with the required minimum rated current (I r ). 4. POWER LINE FILTER WITH TRANSIENT PROTECTION A power line filter of protection class I with a protective conductor is considered below. The circuit diagram in Figure 8 contains all components that are necessary for the filter circuit. The choke L 1 decouples the load, i.e. the power supply, from the mains with its common-mode impedance. This means that the choke is RF-technically with Z1/Z6 and Z2/Z7 when the reference ground (PCB ground to housing) carries interference potential. However, Z 6 and Z 7 are capacitors with a low capacitance (typically 200 - 470 pF) that are inserted to dampen highfrequency harmonics of the switching power supply. The two capacitors should be placed on the circuit board of the power supply if the proportion of switching harmonics on the power supply side is high, or if the reference ground (case ground) does not form a high-frequency stable reference point. The chokes Z 4 and Z 5 are already partially covered by the leakage inductance of the common mode choke L 1 ; Z 4 and Z 5 can also be provided for additional attenuation in the frequency range above 10 MHz. A mains filter for an interface with a 50 Hz mains frequency and a voltage of 230 V AC serves as a calculation example. Figure 8: Circuit diagram of the line filter with transient protection. The maximum permissible leakage current on PE is 3.5 mA according to DIN VDE 0701-0702 for electrical devices with rated voltages up to 1000 V AC /1500 V DC . For safety reasons, a maximum leakage current of I LEAK = 2.6 mA is assumed in the example, from which the maximum capacitance of the Y-capacitors (C Y, MAX ) with the mains filter frequency f = 50 Hz and the worst-case voltage of V MAINS = 257 V is calculated as follows (7) Where: C Y, MAX is the maximum capacitance of the Y-capacitor (F). I LEAK is the maximum leakage current (A). f is the mains frequency (Hz). V MAINS is the possible maximum value of the mains voltage. Substituting the known values in (7), Thus, the maximum possible capacitance value of the Y-capacitor is therefore approximately 33 nF. Step 1: Determination of L 1 When designing the common-mode choke L 1 , the current carrying capacity (rated current Ir) depends on the load, the maximum ambient temperature and the maximum permissible leakage current must be considered. The inductance of the common mode choke is defined depending on the current waveform of the AC-DC converter. For sinusoidal current consumption, a high inductance > 10 mH is used, while for non-sinusoidal current consumption, such as a switching regulator, low inductances in the range of 1 to 10 mH are used. In our example, a sinusoidal current is drawn and a common-mode choke with 10 mH is selected. The current-compensated mains choke WE CMB 744825410 can be used. In the next step, the Y-capacitors Z 1 and Z 2 , and the X-capacitor Z 8 are selected. Step 2: Determination of Z 1 , Z 2 and Z 8 Practical experience for the required minimum attenuation at 150 kHz is 20 dB in differential mode and 40 dB in commonmode. Common values for the filter corner frequencies are 15 kHz in differential mode and 10 kHz in common-mode. The corner frequency represents the frequency at which the attenuation of a filter is 3 dB. Given the required comer frequencies and required attenuation, the capacitances can be calculated: Where: f C , DM is the corner frequency of the differential-mode filter (Hz). f C, CM is the corner frequency of the common-mode filter (Hz). L 1 is the inductance of the common-mode choke (H). L 1,LEAK is the leakage inductance (H), which derived from L1/200 (empirical value). Based on these values, for this example a capacitance of 2.2 µF is chosen for C X and a capacitance of 3 × 4.7 nF = 14.1 nF for C Y . For the Y-capacitors the MLCC Safety Capacitors are used ( WCAP-CSSA Interference Suppression). This results for the attenuation are shown in Figure 9. Figure 9: Differential-mode (green) and common-mode (blue) attenuation by Z 1 , Z 2 and Z 8 . The required attenuation of 20 dB at 150 kHz is exceeded by 32 dB in differential mode. The common-mode attenuation of 40 dB at 150 kHz can also be easily achieved with 43 dB. For transient protection, a disk varistor is used in parallel to the X-capacitor, which is selected next. Step 3: Determination of V R1 There are various approaches to the selection process, such as empirical testing of similar products, selection based on the specific electrical environmental conditions, or the surge test according to IEC/EN 61000-4-5 as a fundamental EMC test. The following section uses the IEC/EN 61000-4-5 standard as the basis. When selecting the disk varistor, the maximum operating voltage (V DC/RMS ) is determined. For safety reasons, this is calculated with a buffer of typically 15% above the supply voltage. With the mains voltage of 230 V AC , the maximum operating voltage is V RMS = 264.5 V AC . As can be seen in Table 4, the size of the varistors depends on the peak current to be clamped. The higher the rated current, the larger the component. Table 4: Ratio of rated current to the diameter of varistors. The disk varistor WE-VD 820443011E with a maximum operating voltage of V RMS = 300 V is used as the basis for the decision. An excerpt from the data sheet can be seen in Table 5. Table 5: Electrical properties of the disk varistor WE-VD 820443011E . According to the IEC/EN 61000-4-5 standard, the voltage of the test pulse for surge between a conductor and a return conductor is V SURGE = 2 kV and the source impedance is 2 Ω for low voltage network connections between live conductor and neutral conductor. The breakdown voltage (V VAR or V BR ) can be found in the varistor datasheet (shown in Table 5). The peak current I PEAK can be calculated as follows: (10) Where: I CLAMP, MAX is the peak current of the varistor (A). V SURGE is the test pulse for surge (V). V BR is the varistor voltage or breakdown voltage at 1 mA (V). Z SOURCE is the source impedance according to IEC/EN 61000-4-5 (Ω). By substituting the known values into (10), the clamped peak current is obtained: To simplify our calculation, we specify that the clamping voltage during the current surge is twice the breakdown voltage VBR. This value is slightly higher than the actual clamping voltage of the varistor at the surge current (refer to the diagram of the rated current vs. voltage in the datasheet). Approximation: V CLAMP ~ 2 V BR . Considering a tolerance of 10%, the maximum clamping current is 583 A and is therefore below the maximum peak current of the varistor (6000 A). Energy consumption and power loss calculation: To further simplify the calculation, it is assumed that the pulse is a rectangular signal that maintains its maximum current value over its 20 µs duration. Then the energy can be calculated with twice the breakdown voltage V BR and I CLAMP, MAX : (11) Where: I CLAMP, MAX is the maximum clamping current. V BR is the varistor breakdown voltage at 1 mA (V). W MAX is the energy consumption of the varistor (J). By substituting the known values into (11), the energy to be absorbed by the varistor is calculated: The IEC/EN 61000-4-5 standard requires at least one surge impulse per minute during testing. The varistor must be able to dissipate this energy. The power requirement can be calculated from the energy and the time between two surge pulses as shown in (12). (12) Where: P is the power dissipated by the varistor (W). W MAX is the energy consumption of the varistor (J). T is the time between two surge pulses (s). Using the known values, the power is calculated according to the (12): The varistor would therefore also be suitable for a surge interval of 20 seconds, whereby the power would then correspond to 549 mW and would therefore be below the maximum power loss of the selected varistor (600 mW). 5. CONCLUSION This support note has explained the basis for transient suppression and basic EMI filtering for both data and power interfaces with worked examples of each that apply the principles discussed. IMPORTANT NOTICE The Application Note is based on our knowledge and experience of typical requirements concerning these areas. It serves as general guidance and should not be construed as a commitment for the suitability for customer applications by Würth Elektronik eiSos GmbH & Co. KG. The information in the Application Note is subject to change without notice. This document and parts thereof must not be reproduced or copied without written permission, and contents thereof must not be imparted to a third party nor be used for any unauthorized purpose. Würth Elektronik eiSos GmbH & Co. KG and its subsidiaries and affiliates (WE) are not liable for application assistance of any kind. Customers may use WE’s assistance and product recommendations for their applications and design. The responsibility for the applicability and use of WE Products in a particular customer design is always solely within the authority of the customer. Due to this fact it is up to the customer to evaluate and investigate, where appropriate, and decide whether the device with the specific product characteristics described in the product specification is valid and suitable for the respective customer application or not. The technical specifications are stated in the current data sheet of the products. Therefore, the customers shall use the data sheets and are cautioned to verify that data sheets are current. The current data sheets can be downloaded at www.we-online.com. Customers shall strictly observe any product-specific notes, cautions and warnings. WE reserves the right to make corrections, modifications, enhancements, improvements, and other changes to its products and services. WE DOES NOT WARRANT OR REPRESENT THAT ANY LICENSE, EITHER EXPRESS OR IMPLIED, IS GRANTED UNDER ANY PATENT RIGHT, COPYRIGHT, MASK WORK RIGHT, OR OTHER INTELLECTUAL PROPERTY RIGHT RELATING TO ANY COMBINATION, MACHINE, OR PROCESS IN WHICH WE PRODUCTS OR SERVICES ARE USED. INFORMATION PUBLISHED BY WE REGARDING THIRD-PARTY PRODUCTS OR SERVICES DOES NOT CONSTITUTE A LICENSE FROM WE TO USE SUCH PRODUCTS OR SERVICES OR A WARRANTY OR ENDORSEMENT THEREOF. WE products are not authorized for use in safety-critical applications, or where a failure of the product is reasonably expected to cause severe personal injury or death. Moreover, WE products are neither designed nor intended for use in areas such as military, aerospace, aviation, nuclear control, submarine, transportation (automotive control, train control, ship control), transportation signal, disaster prevention, medical, public information network etc. Customers shall inform WE about the intent of such usage before design-in stage. 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DIRECT LINK S N024 | Transient suppression at different interfaces USEFUL LINKS: Application Notes : https://we-online.com/en/support/knowledge/application-notes Services: https://we-online.com/en/products/components/service Contact : https://we-online.com/en/support/contact CONTACT INFORMATION Würth Elektronik eiSos GmbH & Co. KG Max-Eyth-Str. 1, 74638 Waldenburg, Germany Tel.: +49 (0) 7942 / 945 – 0 Email: appnotes@we-online.de Web: https://www.we-online.comsupport notewürth elektronikknowledgewurth_elektronikhttps://community.element14.com/products/manufacturers/wuerth-elektronik/b/blog/posts/free-hands-on-training-on-cybersecurityMon, 08 Sep 2025 11:03:00 GMT93d5dcb4-84c2-446f-b2cb-99731719e767:7dfad0e9-5e61-46be-b7a1-1f2ea41c0f03Würth ElektronikTake advantage of the hands-on training with the Cordelia-I EV-Kit on GitHub! This free guide teaches you how to establish a secure wireless connection and implement protocols like MQTT and TLS. Experiment with the EV-Kit and gain practical experience in IoT cybersecurity. Discover now: https://github.com/WurthElektronik/iot-cybersecurity-seminar If you want to know more about Cybersecurity watch our webinars: Cyber Security: Challenges and Standards for the Future Cybersecurity at the Eleventh Hour – from RED to CRA – Information and Discussion The First Step into a Secure Future: Find the Cordelia - I here at Farnell.radio moduleswürth elektronikcybersecuritywurth_elektronikhttps://community.element14.com/products/manufacturers/wuerth-elektronik/b/blog/posts/anp146-theoretical-insights-and-practical-applications-of-the-we-cmdc-series-common-mode-chokesMon, 25 Aug 2025 08:51:00 GMT93d5dcb4-84c2-446f-b2cb-99731719e767:bb2b54dc-62a0-4d48-ab79-1cd54127557cWürth ElektronikApplication Note ANP146 | Theoretical Insights and Practical Applications of the WE-CMDC Series Common Mode Chokes ANP146 by Dr.-Ing. Heinz Zenkner 1. INTRODUCTION Common mode chokes (CMCs) are widely used to filter out interference components in electronic circuits. These components specifically target noise signals that appear on line pairs with the same amplitude and phase relative to the reference ground. Accurately modeling the behavior of CMCs is crucial for understanding the component in detail. Furthermore, the transfer of the model and its functional simulation into practice must be understood in order to achieve their effective integration into complex systems and for enabling targeted optimization of their filtering performance. This paper presents a simulation of a common mode choke and compares the results with real-world measurements. Additionally, it explores unconventional applications of CMCs, expanding the potential use cases for these components. Notably, the parasitic properties of CMCs are also leveraged as functional elements within certain circuit designs, demonstrating their versatility beyond traditional applications. 2. LTSPICE SIMULATION OF CMCs CMCs are passive components that generate a high impedance in a certain frequency range to reduce the propagation of unwanted signals (typically electromagnetic noise). The structure of the CMC is the same as that of a wound coil: There is a core of ferromagnetic material, and several wires wound on this core. Figure 1 shows the basic structure of a CMC. The CMCs in detail differ not only in the number of turns and the design, but also in the type of core chosen for specific applications. The core material has a poor quality factor above a few tens of MHz, i.e. it is chosen so that it causes as many losses as possible above a certain frequency range; Iron powder cores are active at low frequencies (fC 10 MHz up to several hundred MHz). Bifilar and sectional winding styles impact the performance and behavior of the choke, particularly in electromagnetic compatibility (EMC) and filtering applications. Figure 1: Basic structure of a CMC. Left hand side: Bifilar winding style, right hand side: Sectional winding style. In bifilar winding style, two wires (or more) are wound together closely in parallel on the core (left in Figure 1). Bifilar winding provides excellent magnetic coupling between the windings, which enhances the inductance for common-mode currents. Since the wires are wound together, the design is compact and easy to manufacture. The chokes are effective at blocking high-frequency common-mode noise due to the high coupling and resulting inductance. The close coupling cancels the inductance for differential-mode currents, making it less effective at suppressing differential-mode noise. The proximity of the wires increases interwinding capacitance, which can degrade performance at very high frequencies. In sectional winding style (inductor on the right in Figure 1), the windings are separated into different sections on the core. The wires are wound in physically distinct layers or areas, with spacing between them. The separation of windings lowers interwinding capacitance, improving high-frequency performance. Due to less coupling, the choke provides some differential-mode inductance, aiding in suppressing differential-mode noise. The separation can improve heat dissipation. The inductance for common-mode currents is lower compared to bifilar winding. Sectional winding often results in a larger choke, as the windings are spaced out. CMCs are combined with capacitors to form complex filter circuits to achieve the highest possible insertion loss and thus filter effect. How these circuits work, where the weaknesses and strengths lie and how impedance ratios of the system have to be taken into account can best be explained with a simulation, which was carried out below with LTspice. 2.1 Simulation of the CMC WE-CMDC as part of a filter In the following analysis using LTspice, the common mode attenuation and the differential mode attenuation are considered separately. The simulation circuit diagram without choke is shown in Figure 2. The common mode interference signal source (CM) generates a signal with an amplitude of 1 V at a frequency of 5 MHz and is coupled into the circuit via two 200 pF capacitors. The useful signal, the differential mode signal (DM), has an AC voltage amplitude of 1 V at a frequency of 500 kHz and an additional DC voltage amplitude of 10 V, for better visualization in the diagram. Both signals can be viewed at the load R 3 . To illustrate how the CM choke acts as part of a filter, the component is considered under different functional states in the time and frequency domain. The measurements in Figure 3 clearly show that when a single choke (L 3 ) with capacitor (C 10 ) is implemented in one of the two paths, there is no attenuation of the signals with respect to ground, which is the CM signal. Figure 2: Circuit diagram without CM choke. TP 01 to TP 04 show the respective voltages Figure 3: Visualization of the amplitudes of the measuring points in the time and frequency domain (L 3 = 1000 µH, C 9 /C 17 = 100 pF). Both the time and frequency domains are shown; in the time domain, the offset of the signals in the choke path due to the DC voltage can be seen. The time domain shows the 500 kHz and 5 MHz spectral lines at the various measurement points. The levels at the measuring points do not differ in amplitude. Figure 4 shows that only the differential signal is strongly attenuated in its AC voltage amplitude (TP 01 , blue). As the CM interference signal is on both paths, i.e. V+ and V-, both paths must also be provided with filter components to attenuate the interference in relation to ground. For this reason, an unbalanced filter (e.g. LC low-pass filter) cannot be used here. Two chokes were used in the setup in Figure 5. There is a clear attenuation of the interference signal by approx. 30 dB, but the useful signal is also strongly attenuated, as the diagram clearly shows in the time domain. Therefore, the concept of current compensation must be introduced in order not to attenuate the DM signal. In a common-mode choke, the windings act in such a way that the magnetic fields of the currents in differential mode cancel each other out, while they add up in common mode. The current in differential mode flows through the windings in opposite directions, whereby the magnetic fields generated ideally compensate each other completely. As a result, the inductance for differential currents approaches zero, which does not attenuate the normal signal flow. With common mode currents, the currents flow in the same direction through both windings, whereby the magnetic fields are added together, and a high inductance is created. Figure 4: Visualization of the amplitudes of the measuring points behind the filter in the time domain, (L 3 = 1000 µH, C 9 /C 17 = 100 pF). Figure 5: Filter with two chokes (L 1 , L 2 ), Both DM and CM signals are attenuated. TP 05 /TP 06 show the differential-mode attenuation (signal at 500 kHz), TP 01 /TP 02 and TP 03 /TP 04 show the common mode attenuation (“noise” at 5 MHz). T his high inductance provides a high impedance for high-frequency common mode interference and effectively attenuates it. The decisive factor for high common mode attenuation is a high inductive coupling of the two coils, the so-called mutual inductance. This is the phenomenon whereby a change in current in one circuit induces a voltage in a neighboring circuit due to the magnetic field generated by the current, as in a transformer. The degree of coupling between the two circuits is quantified by the mutual inductance M, which depends on factors such as the number of turns in the coils, their relative orientation/position, the distance between them and the magnetic properties of the medium. The mutual coupling factor K quantifies the coupling between the two windings of the choke. It is defined as: (1) where: M is the mutual inductance between the windings. L 1 and L 2 are the self-inductances of the windings. K ranges from 0 (no coupling) to 1 (perfect coupling); if the coupling factor K is 1, the mutual coupling of the two inductors is 100%. Typical coupling factors for common mode chokes are between 0.80 and 0.95. The effect of the mutual inductance was used in Figure 6. It clearly shows that the useful signal at 500 kHz is no longer attenuated, but the CM interference signal at 5 MHz is reduced by approx. 40 dB However, the attenuation in the current system is strongly dependent on the load impedance. To achieve sufficient attenuation even for high-impedance CM loads, such as peripheral cables (→ monopole antenna with high impedance in the event of mismatch), the output of the filter must be low impedance in terms of RF technology. For this reason, additional capacitors, so called Y-capacitors, are implemented at the output of the CM choke. This results in a frequency-dependent voltage divider, as shown in Figure 7. As now the filter is a frequency-dependent voltage divider, the divider principle makes the filter independent of the load impedance. The set up in Figure 7 shows that the DM signal is hardly attenuated at all, whereas the CM signal is reduced by over 70 dB due to the addition of the Y-capacitors. As can be seen, the implementation of the Y-capacitors has only a minor effect on the attenuation of the wanted signal, as the value of the capacitor (here 100 nF) only loads the wanted signal with half its capacitance. Of course, it must be mentioned here that, firstly, the useful and interference signals are far apart in terms of frequency and that no differential-mode inductance is implemented here, which is explained next. As mentioned before, the mutual coupling factor K quantifies the coupling between the two windings of the choke. On the other hand, the mutual coupling factor K of a CM choke not only indicates the strength of coupling between windings, but also reflects the level of stray inductance. A special “trick” can be used for additional differential mode attenuation. When the mutual coupling factor is reduced, a leakage inductance (stray inductance) is created which can be used as part of the DM inductance and thus as a DM filter; again, with an additional capacitor (X-capacitor) connected in parallel to the load. In many cases, the Y-capacitors, which add up to half their value as an X-capacitor, are already sufficient. A high K means strong coupling and minimal stray inductance, which reduces the choke’s impedance to DM currents, lowering its DM attenuation. A lower K corresponds to higher stray inductance, increasing DM impedance and improving DM noise suppression. However, a high K is critical for maximizing CM attenuation, as it increases the choke’s impedance to CM currents. The design must balance K to manage the trade-off between CM and DM attenuation or use additional components to address DM noise such as SMD ferrite beads, to increase the attenuation in high frequency range, but to minimize the influence of attenuation on useful signals in the lower frequency range. However, the statement is fully applicable to sectional winding, where K can be adjusted to balance CM and DM attenuation. For bifilar winding, K is inherently high, so the trade-off does not apply in the same way. Instead, additional measures like ferrite beads are needed for DM attenuation. Figure 6: Filter with two inductors in mutual inductive coupling. The signals of TP 01 and TP 02 overlap in time domain, the signal of TP 02 is behind the signal of TP 01 in the graph. Figure 7: Common mode choke filter with additional Y-capacitors (C 22 , C 23 ). The signals of TP 03 and TP 04 overlap in time domain, the signal of TP 03 is behind the signal of TP 04 in the graph. Figure 8 shows the influence of the coupling factor K on the attenuation of the DM attenuation, for better visualization the CM source level was set to 0. Figure 8 shows that even a small reduction in the K-factor can have a major effect on the attenuation of the differential signal. In the present simulation, an inductor without the non-linear behavior of the ferrite material on the impedance over the frequency is used. In practice, however, the course of the complex permeability over the frequency and thus the non-linearity of the inductance over the frequency must be taken into account. Furthermore, the parasitic parameters, such as the parasitic capacitances of the windings, must be considered. The first step in acquiring these parameters is to determine the equivalent circuit diagram of the CM choke Figure 8: Influence of the K-factor to the DM attenuation. 2.2 Measurement and calculation of the parameters for the equivalent circuit diagram of the CMC. The equivalent circuit diagram of the choke is required not only for the simulation of the CMC but also for understanding the component if the filter insertion loss is to be considered taking into account the real properties. Data can be taken from the data sheet, but some parameters must also be measured and calculated. The following is a summary. When developing the equivalent circuit diagram of a CMC, the parameters of the equivalent circuit diagram are mainly determined by measurements, which are carried out in three steps. Figure 9 shows the equivalent circuit diagram used here. The selected common-mode choke is the WE-CMDC, Order Code: 744238132 , the data sheet shows the following parameters that can be used to further determine the missing values (Figure 10) Figure 9: The equivalent circuit diagram of the CM choke used here. The equivalent loss resistance (R LOSS_X ), in has been converted here from a parallel circuit to a series circuit Figure 10: Electrical parameters from data sheet, CM choke WE-CMDC, 744238132 The choke has an inductance per winding of 10.40 µH, the inductance was measured at a frequency of 100 MHz. At this frequency, a scalar impedance in the range of 910 – 1300 Ω is specified. The DC resistance, i.e. the copper resistance of a winding, is a maximum of 25 mΩ. The blue curve in the upper diagram in Figure 10 shows the CM impedance, which corresponds to the impedance of a winding (L 1 or L 2 ) with the ferrite core, as the windings are connected in parallel in the same winding direction. The maximum of the curve (dark blue) is at approx. 100 MHz, the resonance is not very pronounced. A direct linear increase in impedance from 1 MHz is not recognizable, i.e. the ferrite material has a lossy permeable part over the entire frequency range, which leads to the flat curved shape of the curve. The lower diagram in Figure 10 shows the DM impedance. Up to approx. 50 MHz the impedance increase is linear. The diagram also shows a resonant frequency at approx. 180 MHz. The resonant frequency is pronounced, it is caused by the parasitic coupling capacitances of the windings. In the next step, the impedance of the primary coil is measured, and the ferrite loss resistance and the parasitic winding capacitance are determined. The basic measurement setup and the corresponding equivalent circuit diagram are shown in Figure 11. Figure 11: Schematic measurement setup for determining the parameters for the equivalent circuit diagram of the CM choke. The inductance L 0 , including the leakage inductance L S , the resistance R 0 , which corresponds to the ferrite loss, and the leakage capacitance C 0 of the winding are measured. R 0 and C 0 are the total values of the primary and secondary coils. L 0 is the sum of the magnetic fluxes generated by the primary and secondary coils. The inductance appears to double, but the primary and secondary coils are connected in parallel, so the result remains L 0 . The result of the impedance measurement is shown in Figure 12. To determine the inductance, an area of the impedance curve is selected, that is as linear as possible. Figure 12: CM impedance of the coil in the frequency range between 100 Hz and 50 MHz, red: impedance, blue: phase. In this area, the phase shift of the impedance should be over 80°, as this indicates a dominant inductance and capacitive and ohmic losses are negligible. This ensures that the inductance is calculated based on a realistic model. If the inductance were calculated in a range with significant loss components (phase components below 80°), larger deviations could occur, as the loss resistance falsifies the measurement. The diagram in Figure 12 shows that at 100 kHz the impedance is approx. 12 Ω at a phase angle of 82°. This results in the following calculation: Z at 100 kHz: 13 , θ = 82° With results in: Resistance R: 1.67 Ω → R O = 3.34 Ω Inductive Reactance X L : 11.88 Ω Inductance L O : 18.91 µH To calculate the leakage inductance, the impedance of the primary coil can be measured with the secondary coil short-circuited. Due to the magnetic coupling, L S is the sum of the leakage inductances of the primary and secondary coils. If the secondary coil is short-circuited, the magnetically coupled L 0 is short-circuited from the primary coil's point of view and the residual leakage inductance L S can be measured. The schematic measurement setup is shown in Figure 13. L 0 is the total inductance, consisting of leakage inductance and the inductance L 0 ' , which acts on the common mode interference. This results in the following formula: (5) Figure 14 shows the measurement protocol for calculating the leakage inductance. The following calculations are obtained with Z = 1.3 Ω and θ = 76˚ at 100 kHz: Resistance (R) is calculated using R = Z · cos(θ) Inductive Reactance (X L ) is calculated using: X L = Z · sin(θ) Inductance (L) is related to X L by: L = X L /(2 · π · f) Figure 13: Schematic measurement setup for determining the leakage inductance. Figure 14: Measurement protocol for calculating the leakage inductance. Red: Impedance, blue: Phase. (9) Results: Resistance (R): 0.31 Ω Inductive Reactance (X L ): 1.26 Ω Inductance (L S ): 2.01 µH, which is the stray or leakage inductance, measured at 100 kHz. The parasitic capacitance is determined at the resonance point of the choke, which is above the frequency of 50 MHz. For this purpose, the insertion loss is measured as shown in Figure 15 using an RF network analyzer. Three different signal levels were measured in order to take into account any shifts caused by the influence of the ferrite material. The curves in Figure 15 all show a resonance at 241.77 MHz. The stray capacitance can now be calculated as follows. With (10) and (11) and C 0 = 2 · C S ; L S = 2 µH There is: C 0 = 1.26 pF (including all measurement stray effects as well). The next, very important parameter of a CM choke is the coupling coefficient K or the mutual inductance M. As mentioned before, the coupling coefficient K of a CM choke is a measure of how effectively the magnetic fields of its two windings are coupled. Ideally, for a CM choke, K should be close to 1, meaning the magnetic flux generated by one winding is almost entirely linked with the other winding. This high coupling ensures effective suppression of CM noise while allowing differential signals to pass with minimal interference. But in this case as there will be no stray inductance for filtering DM signal harmonics or noise, an additional inductance must be added. Thus, if the coupling coefficient K is reduced the stray inductance might be useful to reduce the DM noise which of course then must be in the frequency range above the useful signal. Figure 15: Insertion loss of the DM impedance with three different signal/measurement levels. The following relationship exists for the parameters K and M: Coupling coefficient K = (L 0 – L S )/L 0 Mutual inductance M = K L 0 L 0 = 18.91 µH L S = 2.01 µH This results in the following values: Coupling coefficient K = 0.89 Mutual inductance M = 16.9 µH The measurement of the copper resistance R DC at direct current is still missing, which is surprisingly difficult, as this measurement of a very small resistance value depends on the ambient temperature and the test object contact (despite Kelvin terminals). Ultimately, the value was measured under controlled conditions; the setup and measurement results are shown in Figure 16. As the primary and secondary coils have the same number of turns, the measured value also applies to the secondary coil. The value determined for the copper resistance R DC of a winding including the connections is 23 mΩ. Figure 16: Schematic measurement setup for determining the copper resistance of the windings. Overall, this results in the following equivalent circuit diagram for the CM inductance (Figure 17). Figure 17: Equivalent circuit diagram of the CM choke WE-CMDC, order code: 744238132 . The influence of the parameters C 0 and K on the filter properties are shown below. The circuit diagram shown in Figure 18 is used to show the influence of the parasitic capacitance C 0 of the CMC Figure 18: Circuit diagram and output voltage (V OUT ) of the filter to show the influence of C 0 . As this is a simulation, no capacitive stray effects due to measurement set-up are included. The parasitic capacitance in the windings of this inductor arises from the proximity of adjacent turns of the windings, leading to unintended capacitive coupling. This coupling forms a parallel resonance with the inductance of the windings, creating a self-resonant frequency which is clearly visible in Figure 18. Below the self-resonant frequency, the inductor exhibits primarily inductive behavior, providing effective common-mode noise attenuation. However, as the frequency approaches the self-resonant frequency, the parasitic capacitance causes the inductor's impedance to drop significantly. At frequencies above the self-resonant frequency, the inductor's impedance becomes capacitive, reducing its effectiveness in attenuating CM noise. This shift adversely affects the insertion loss, particularly at higher frequencies, where the inductor no longer provides significant attenuation. As can be seen in Figure 18, even small values of parasitic capacitance such as 0.3 pF lead to strong resonance effects at a frequency in the range of around 115 MHz. This pronounced parallel resonance has a high Q factor. For this reason, the presence of parasitic capacitances can introduce unwanted resonances into the circuit, potentially amplifying certain noise frequencies instead of suppressing them, and easily coupling into neighboring circuits. To mitigate these effects, careful design practices are used, such as selecting the proper inductor, minimizing coupling at the PCB, damping resonances with additional losses like resistors, and adding differential mode SMD ferrites that are lossy in this frequency range. These measures help minimize parasitic capacitance and extend the effective operating frequency range of the inductor. The circuit diagram shown in Figure 19 is used to show the influence of the coupling factor K, resp. the Mutual inductance M of the CMC over the DM voltage V OUT . A high coupling factor, close to 1, indicates strong magnetic coupling, meaning that most of the flux generated by one winding links with the other. This enhances the inductance seen by CM currents, improving the inductor's ability to attenuate CM noise. Consequently, the resonant frequency for CM noise is lowered due to the increased effective inductance. The resonant frequency is primarily influenced by the leakage inductance and the parasitic capacitance. Thus, as shown in Figure 19, a higher coupling factor reduces leakage inductance, which in turn raises the resonant frequency. With a higher K, the inductor provides better CM noise attenuation across a broader frequency range before reaching resonance. Additionally, the insertion loss for CM noise improves because the inductor remains effective at higher frequencies. Conversely, if K is low, the leakage inductance increases, lowering the resonant frequency and potentially reducing the effective frequency range for noise suppression. This may also result in a lower insertion loss at higher frequencies. In practical applications, a CMC needs to be selected which balances high coupling for effective noise suppression with a proper selected leakage inductance to prevent unintended impacts on DM signals Figure 19: Circuit diagram and output voltage (V OUT ) of the filter to show the influence of M (K). 3. FILTER APPLICATIONS The following section presents practical examples that demonstrate the use of CM chokes. These applications are straightforward and highlight the benefits of CM chokes when their design and parameter selection are carefully optimized. 3.1 Common Mode Filter Figure 20 shows the schematic of a simple CM filter. To show the effects of selecting different values of the WE-CMDC series the insertion loss was measured. The filter is built on a breadboard with a GND layer; the GND layer was omitted in the area of the CM choke in order to keep the capacitive coupling small. The input signal is a differential signal with an impedance of 50 Ω to ground, i.e. 100 Ω differential impedance. The output was fed to the oscilloscope via a differential probe with 1 MΩ differential input impedance. Figure 21 shows the DM attenuation. The double logarithmic diagram in Figure 21 shows that the differential attenuation is low for all selected values up to approx. 5 MHz. Beyond this frequency rage, the attenuation depends not only on the selected inductance value, but also on the behavior of the imaginary part of the permeability µ'' of the ferrite material. Of course, the filter with the higher inductance also shows a higher attenuation, but the non-linearity, especially the kink at approx. 35 MHz, is due to the ferrite material. Above 35 MHz, the attenuation increases significantly for all 4 filters, as the ferrite material comes closer to its resonant frequency and thus its higher impedance range where not only µ', but also µ'' increases significantly. Figure 20: Schematic and setup of a CM filter, using different common mode chokes of the WE-CMDC series. “IN” is the side on which the measurement signal (in practice the interference signal) was applied. Figure 21: DM attenuation of the filter, shown in Figure 20. Mutual inductance (M) decreases with frequency as the ferrite’s real permeability (µ') drops, increasing L STRAY and DM attenuation. The ferrite’s imaginary permeability (µ'') increases at higher frequencies, introducing resistive losses that further enhance DM attenuation. Although L STRAY behaves like an air-core inductor, its value is influenced by the frequency-dependent coupling (M) with the core material. As a result, DM attenuation depends on the ferrite material because its permeability variations affect both L STRAY and resistive losses across the frequency range. The CM attenuation is shown in Figure 22. A higher inductance leads to a higher impedance, the markers each show the 3 dB cut-off frequency. All chokes show no pronounced resonances, in the range around 80 MHz there is a weak parallel resonance due to the design. In the range between 200 MHz and 300 MHz the impedance turns the chokes into the capacitive range, the impedance increases. This range is also determined by the measurement assembly and can vary depending on the layout and setup. Nevertheless, in the 50 Ω system of the network analyzer, with all inductance values, the setup enables at least an insertion loss of 40 dB at 100 MHz, with the 22.6 µH choke over 50 dB. 3.2 Coupled Filter for noise reduction in DC-DC converters Noise on the outputs of DC-DC converters can originate from several sources, including high-frequency switching transients, conducted and radiated electromagnetic interference (EMI), and ripple voltage due to imperfect filtering. The switching process in pulse-width modulation or pulse-frequency modulation converters generates high-frequency noise that can couple into sensitive circuits. Insufficient filtering or improper PCB layout can exacerbate these noise issues, leading to signal integrity problems. When this noise propagates into circuits, it can cause malfunctioning of analog and digital components, leading to instability, erroneous data processing, or degraded performance. High ripple voltage can interfere with precision analog circuits, introducing offset and reducing accuracy in sensor readings or ADC conversions. In RF systems, noise from DC-DC converters can increase phase noise, reducing overall system sensitivity. For high-speed digital systems, excessive noise can result in timing errors and communication failures. Figure 22: CM attenuation of different chokes in the range from 500 kHz to 1 GHz, measured in the 50 Ω-system with a network analyze. An additional filter, which is constructed from a CM choke in a special switching mode, can effectively reduce residual interference, Figure 23 shows the circuit diagram. Figure 23: DC-Line filter for additional noise attenuation with WE-CMDC as a coupled inductor. “V IN ” is the side on which the measurement signal (in practice the interference signal) was applied. By coupling the second winding to ground, an additional reduction in noise can be achieved. One parameter for a high attenuation is of course the inductance of L 1 , another is the mutual coupling factor, which should be in the range below 0.9 as the stray inductance determines the attenuation. The higher the capacitance of C 1 , the lower the frequency range that can be attenuated. A typical value is 220 µF, which can be used to filter into the kHz range. The ESR of the capacitor should be low, which is why a polymer capacitor was used. Some measurement signals to illustrate the effectiveness are shown in Figure 24. The signal attenuation at 1 MHz is approx. 73 dB; if capacitor C 1 is removed, the attenuation is reduced by approx. 6 dB. This reduction in attenuation is practically constant from 200 kHz up to 30 MHz. Figure 24: Signal attenuation and insertion loss of the filter with coupled inductor. 3.3 Oscillator with very low distortion If you turn the CM inductor by 90 degrees in the circuit, you get a signal transformer. Of course, the CM choke was not manufactured for this application, but the “usefulness” of the component for this application can easily be checked. Figure 25 shows the transfer characteristic of the 22.6 µH choke over frequency. Of course, the signal transformer can only be used in a frequency range in which the transmission characteristics are good, i.e. the linearity is high, and the losses are lowest. According to Figure 25, this is the range between approx. 80 kHz and 2 MHz. The following application shows the use of the CMC choke as a signal transformer in a Meissen oscillator. The Meissen oscillator circuit generates sinusoidal oscillation. It consists of an amplifier (e.g. a transistor) and a signal transformer (two coupled coils), which serve as a frequency-determining element. The positive feedback to generate oscillation is realized by the inductive coupling between the coils of the transformer. The oscillation frequency is determined by the inductance of the transformer and the capacitances in the circuit. The oscillator is easy to set up and offers a stable oscillation generation with a high output amplitude. The frequency can be easily adjusted by changing the inductance or capacitance. As the output at C 2 is not decoupled, any capacitive load at C 2 changes the resonant frequency. R 3 decouples the resonant circuit and thus mitigates the effects of capacitive loads on the resonant frequency. Figure 26 shows the schematic, the set-up and the output signal of the circuit. Figure 25: Transfer characteristic of the CM choke if used as a signal transformer with the turn ratio 1:1 (N a :N b ) Figure 26: Schematic and the output signal of the Meissen oscillator. With a supply voltage of 15 V, the output voltage is 10 Vpp into a 1 MΩ oscilloscope probe. The output signal has a frequency of 1 MHz, and the spectrogram shows that the harmonics of the signal are in the noise, thus having a distance to the fundamental signal of more than 60 dB. This makes the circuit very interesting for applications where high signal purity is required. 3.4 Primary DC-filter, common- and differential mode The last application in this article is the use of WE-CMDC in a DC input filter, i.e. the classic common mode choke. However, transient and reverse polarity protection are also taken into account in this application (Figure 27). What is important here is a low-offset-interference, low-impedance connection to the primary ground (SGND) for the effectiveness of the "Y-capacitors" to effectively attenuate the CM interference. It is also important to ensure that the transition from SGND to GND on the secondary side of the CM choke is done at a controlled connection on the metal housing. For a detailed description, please refer to the reference design RD041 . 4. CONCLUSION It has been shown that a CMC choke with a ferrite core is an effective measure for suppressing CM interference in electrical and electronic circuits. However, it is important to understand that inductance is not the only parameter that needs to be considered in terms of signal characteristics. The frequency-dependent permeability of the ferrite material and the parasitic capacitance of the choke change the characteristics. Careful selection of the choke, taking into account the frequency of use and the specific application, is therefore crucial. The application areas of a CMC may well go beyond the typical areas such as EMC filters in power supply units. Nevertheless, optimal design requires consideration of all electrical and mechanical properties to ensure effective function. Figure 27: Schematics of a DC-filter with CM and DM attenuation. A APPENDIX Finally, Figure 28 in the Appendix provides a comprehensive overview of typical inductors used in signal applications. The focus is primarily on common-mode chokes, complemented by SMD ferrites and hinged ferrite series. This comparison aims to help the user better understand the key differences, advantages, and specific applications of each component, enabling more informed and effective selection. Figure 28: Overview of typical Inductors for Signal Applications: Common-Mode Chokes, SMD Ferrites, and Hinged Ferrite Series IMPORTANT NOTICE The Application Note is based on our knowledge and experience of typical requirements concerning these areas. It serves as general guidance and should not be construed as a commitment for the suitability for customer applications by Würth Elektronik eiSos GmbH & Co. KG. The information in the Application Note is subject to change without notice. This document and parts thereof must not be reproduced or copied without written permission, and contents thereof must not be imparted to a third party nor be used for any unauthorized purpose. Würth Elektronik eiSos GmbH & Co. KG and its subsidiaries and affiliates (WE) are not liable for application assistance of any kind. Customers may use WE’s assistance and product recommendations for their applications and design. The responsibility for the applicability and use of WE Products in a particular customer design is always solely within the authority of the customer. Due to this fact it is up to the customer to evaluate and investigate, where appropriate, and decide whether the device with the specific product characteristics described in the product specification is valid and suitable for the respective customer application or not. The technical specifications are stated in the current data sheet of the products. Therefore the customers shall use the data sheets and are cautioned to verify that data sheets are current. The current data sheets can be downloaded at www.we-online.com. Customers shall strictly observe any product-specific notes, cautions and warnings. WE reserves the right to make corrections, modifications, enhancements, improvements, and other changes to its products and services. WE DOES NOT WARRANT OR REPRESENT THAT ANY LICENSE, EITHER EXPRESS OR IMPLIED, IS GRANTED UNDER ANY PATENT RIGHT, COPYRIGHT, MASK WORK RIGHT, OR OTHER INTELLECTUAL PROPERTY RIGHT RELATING TO ANY COMBINATION, MACHINE, OR PROCESS IN WHICH WE PRODUCTS OR SERVICES ARE USED. INFORMATION PUBLISHED BY WE REGARDING THIRD-PARTY PRODUCTS OR SERVICES DOES NOT CONSTITUTE A LICENSE FROM WE TO USE SUCH PRODUCTS OR SERVICES OR A WARRANTY OR ENDORSEMENT THEREOF. WE products are not authorized for use in safety-critical applications, or where a failure of the product is reasonably expected to cause severe personal injury or death. Moreover, WE products are neither designed nor intended for use in areas such as military, aerospace, aviation, nuclear control, submarine, transportation (automotive control, train control, ship control), transportation signal, disaster prevention, medical, public information network etc. Customers shall inform WE about the intent of such usage before design-in stage. In certain customer applications requiring a very high level of safety and in which the malfunction or failure of an electronic component could endanger human life or health, customers must ensure that they have all necessary expertise in the safety and regulatory ramifications of their applications. Customers acknowledge and agree that they are solely responsible for all legal, regulatory and safety-related requirements concerning their products and any use of WE products in such safety-critical applications, notwithstanding any applicationsrelated information or support that may be provided by WE. CUSTOMERS SHALL INDEMNIFY WE AGAINST ANY DAMAGES ARISING OUT OF THE USE OF WE PRODUCTS IN SUCH SAFETYCRITICAL APPLICATION. DIRECT LINK ANP146 | Theoretical Insights and Practical Applications of the WE-CMDC Series Common Mode Chokes USEFUL LINKS: Application Notes : https://we-online.com/en/support/knowledge/application-notes Services: https://we-online.com/en/products/components/service Contact : https://we-online.com/en/support/contact CONTACT INFORMATION Würth Elektronik eiSos GmbH & Co. KG Max-Eyth-Str. 1, 74638 Waldenburg, Germany Tel.: +49 (0) 7942 / 945 – 0 Email: appnotes@we-online.de Web: https://www.we-online.comwürth elektronikapplication_noteknowledgewurth_elektronikhttps://community.element14.com/products/manufacturers/wuerth-elektronik/b/blog/posts/ans018-emi-filter-design-for-non-isolated-dc-dc-convertersSun, 17 Aug 2025 07:35:00 GMT93d5dcb4-84c2-446f-b2cb-99731719e767:5087129d-a33b-44d5-bc5d-2f0212383cbfWürth ElektronikApplication Note EMI Filter Design for Non-Isolated DC/DC Converters ANS018 by Timur Uludag 1. INTRODUCTION Today, many industrial applications run with input voltage levels such as 5V DC or lower, as shown in Figure 1. The power distribution system used to power such applications is often a DC bus voltage of 24V DC . DC/DC power modules are commonly chosen to handle the conversion of the higher DC bus voltage down to the lower voltage level. A clear advantage that comes with the use of power modules is that very high efficiencies (often greater than 90%) can be realized. However, the switching behavior of these modules present additional design challenges. Constant switching during operation generates interference energy that can a negative impact on the components in the application. Therefore, developers should always check whether EMI filtering measures are required in their design when using DC/DC power modules. Figure 1 shows the block diagram of a typical industrial environment application with a 24V DC bus. Non-isolated power modules are used to provide the operating voltage of all subsystems. This indicates the importance of DC/DC converters for industrial applications, since these existing subsystems must be supplied with different voltage levels. The converted voltages can be used to supply programmable logic controller systems or further peripheral interfaces for data acquisition, data transmission or human-machine interfaces such as industrial control panels. The DC/DC power module 171032401 is a fully integrated DC/DC converter including the switching regulator IC with integrated MOSFETs, compensation and a shielded inductor in a single housing. For physical reasons, the switching processes of the MOSFETs cause EMC interference in every DC/DC converter, whether discrete or fully integrated as in the power module. Based on the switching frequency, this interference can occur in the EMC measurement, e.g. as conducted harmonic interference amplitudes. Therefore, a fundamental understanding of the EMI basics is essential for the topic of EMI filter for Non-Isolated DC/DC Converters. Chapter 2 and 3 will provide the basics about the types of emissions and the noise modes here. Figure 1: Typical industrial application environment. 2. TYPES OF EMI EMISSIONS 2.1 Conducted EMI Conducted emissions are interferences that propagate via the connecting lines of the device. This type of interference voltage is usually specified logarithmically as dBµV and is an alternating voltage. Figure 2: Block diagram of conducted EMI. The arrows in Figure 2 symbolize disturbances that flow in different directions from different systems and can therefore be both sources and sinks. In case of power modules, the IEC 55032 / CISPR32 standard "Electromagnetic compatibility of multimedia equipment - Emission Requirements " (derived from IEC 55022 / CISPR22 and CISPR13) is used as a reference for the interference voltage limit values. This standard is used in the field of information technology. It contains the limits for the interference voltage in the frequency range from 150 kHz to 30 MHz. Two classes are defined regarding the application area: Class A: For use in industrial environment. Class B: For use in the immediate vicinity of residential, business and commercial areas. The noise levels are divided into Max Peak, Quasi-peak and Average. The Max Peak only indicates the peak value of the interference level. The quasi-peak value evaluates the “modulation frequency” and intensity, i.e. amplitude of a signal. The more frequent the signal appears in its periodicity and the "louder" it is, the higher the quasi-peak value. This is used to evaluate pulsed signals with a low repetition rate. Figure 3 shows the CISPR32 limits for the Quasi-peak and average levels. Figure 3: Limits for conducted EMI according to IEC 55032 / CISPR32. 2.2 Radiated EMI In contrast to conducted interference emissions, radiated interference emissions are the transmission of interference from the interference source to the interference sink across the space and are measured in dBµV/m. Figure 4 represents the radiation of the power module and the connected load. Figure 4: Schematic of radiated EMI. The limits for the radiated interference field strength in the frequency range from 30MHz to 1GHz can also be taken from the CISPR32 standard. These are shown in Figure 5. Figure 5: Limits of radiated EMI according to IEC 55032 / CISPR32. 3. NOISE MODES 3.1 Differential Mode In the case of differential mode (DM) interference, the interference current flows in opposite directions in the forward and return conductors. These currents cause a corresponding voltage drop at the impedance of the interference sink. Figure 6 illustrates the DM current I DM between the interference source and the interference sink using a non-isolated MagI³C Power Module with a load connected at the output as an example. Figure 6: Differential mode interference. 3.2 Common Mode The interference current I CM of common mode noise (CM) flows in the same direction in the forward and return conductors. In a symmetrical circuit design, these currents do not cause a voltage drop at the interference sink between the forward and return conductors. However, the interference voltage V CM can be measured against ground. In addition to the DM input current, a common-mode current is most frequently observed with isolated power modules. The common mode circuit is closed by the coupling capacitances between primary and secondary windings. To enable current flow in the same direction on the forward and return path, the stray capacitances to ground must be taken into account. Consideration of these stray capacitances is the only way for the noise circuit to be closed. This is shown in Figure 7 using an isolated MagI³C Power Module with a connected load as an example. The stray capacitances are symbolized as C P1 with connection to ground. Figure 7: Common mode interference. 4. REDUCE INTERFERENCE THROUGH PASSIVE FILTERS Passive filters are frequency dependent current and voltage dividers made of L, C and R components. These current and voltage dividers reduce spurious emissions from the power module at the interference sink. However, the filtering effect in real passive devices is limited because they always have parasitic elements. For example, above the resonant frequency, the capacitor increasingly behaves like an inductor and exhibits inductive phase behavior. This means that the phase shift gradually changes from -90° (capacitive) to +90° (inductive). The capacitor now acts as an inductive impedance, as the parasitic inductance dominates, and the impedance increases with increasing frequency. Similarly, the capacitive influence increases with inductors above the resonant frequency. The phase shift gradually moves into the negative range and approaches -90° (capacitive behavior), whereby the progression is continuous and dependent on the frequency. In this range, the coil is more or less a capacitor, as the parasitic capacitance now has a stronger effect than the inductance. However, this does not mean that the components no longer have any filter effect from this point on. Instead, it means that from this frequency on, careful attention must be paid to the impedance of the component to determine the corresponding filter effect. But since this application note is about conducted interference in the frequency range up to max. 40MHz regarding DC-DC converters most passive components (inductors & capacitors) can be used for EMI suppression. The larger challenge with the circuits considered is to find a clean reference ground for the capacitor and to prevent "over coupling" of the filter. 1 1 This application note focuses on passive filters for non-isolated MagI³C power modules and thus on filtering of DM interferences. For an understanding of the filtering of CM interferences as well as the passive filter design for isolated MagI³C power modules, at this point reference is made to the WE application note ANS022 . 5. EMI FILTER DESIGN FOR NON-ISOLATED MAGI³C POWER MODULES In order to effectively filter both conducted interference and radiated interference from the power module, a series impedance (choke) and a bypass impedance (capacitor) are added to the input circuit, resulting in a low-pass filter with an attenuation of 40 dB/decade. Normally, the power module is operated with a necessary input capacitor, which must be considered in the filter configuration. The resulting filter design is generally referred to as a PI filter. Figure 8 shows the circuit diagram of the LC filter including the interference sink (LISN) 2 and a non-isolated MagI³C power module shown as the interference source (constant current source). Here the complex impedance Z BP1 corresponds to the input capacitor, Z BP2 corresponds to the added bypass capacitor, Z L to the series inductance and Z V to the impedance of the interference sink. 2 Here only conducted emissions up to 30MHz are considered. Above 30 MHz the impedance concept of the LISN doesn’t work anymore because it is supposed to replicate the typical impedance of a household mains connection. Figure 8: EMC model with the MagI³C power module as current source. To simplify the model, the constant current source is converted into a constant voltage source, whereby the first bypass impedance Z BP1 (impedance of the input capacitor) is integrated into the voltage source. This can be done by considering the input capacitor as a very low impedance compared to the inductor impedance and the LISN impedance. This means that the input capacitor is the dominant part and defines the voltage by drawing the main current in this setup. This results in an LC filter due to the series impedance, Z L , and the second bypass impedance, Z BP2 . The resulting circuit diagram is shown in Figure 9. Figure 9: EMC model with the MagI³C module as voltage source. Using the assumptions just discussed, the attenuation of an LC filter can be calculated as follows. The voltage V IN is the voltage that occurs when the power module is connected only with input capacitors to the LISN. Consequently, A 1 corresponds to the measured amplitude of a power module without further filtering and A 2 corresponds to the measured amplitude with an additional filter. The difference between A 2 and A 1 is the damping D. This formula can be simplified if the dominant parts of the circuit are considered. It can therefore be assumed that if a capacitor is used as Z BP2 , this is the dominant part and the parallel circuit can be reduced to this. This results in the following formula: This formula can be further simplified under the condition that Z L ≫ Z BP2 . This is due to the fact that Z L is the dominant part for the current in this curcuit. To determine the component values, this formula must be converted to use C and L values, see Figure 10. When designing passive filters, it must be considered that the filter components are not ideal and have parasitic impedances as mentioned already. Furthermore, the components often show a derating with respect to voltage and/or current. For example, the capacitance of a capacitor, depending on its material and design, decreases with an increasing DC bias voltage, likewise an inductor has a lower inductance with a higher DC current and must not reach core saturation (Reference Guide: Trilogy of Magnetics). 6. INPUT EMI FILTER FOR MAGI³C POWER MODULES From the previous chapters, the filter design was explained theoretically. For the practical implementation, however, the exact components recommended will have an important role to play in creating an effective filter. The online design tool Redexpert4 from Würth Elektronik is perfectly suited for this purpose, as it maps the impedance curves of all the filter components. Additionally, use its EMI Filter Designer to recommend values and plot the resulting response. Due to the frequency behavior of MLCCs as well as polymer capacitors, these capacitor types are very well suited to be filter capacitors. The WE-PD2 series of inductors offers excellent characteristics for the role of filter inductance. To support developers in circuit design, Würth Elektronik offers application-specific filter configurations for isolated and non-isolated power modules. Figure 10 shows an example of a filter circuit of a non-isolated MagI³C power module with variable output voltage using the TO-263EP package ( VDRM ). The module requires additional, external input capacitance (C 2 and C 3 ) to optimize the input current ripple. This input capacitance should be placed as close as possible to the power module. Figure 10: Input EMI filter for non-isolated MagI³C Power Modules. This approach to input filter design is valid for the entire non-isolated MagI³C power module portfolio. Figure 11 and Figure 12 show the measured conducted emissions according to the IEC55032 / CISPR32 test setup with the evaluation board 178032401 of the 171032401 MagI³C module to provide real validation of the filter design. Figure 11: Measured conducted emission of MagI³C VDRM 171032401 without input filter. Figure 12: Measured conducted emission of MagI³C VDRM 171032401 with input filter. The diagram shows that the limit values of the IEC55032 / CISPR32 are exceeded at the switching frequency of 500 kHz as well as with some of the following harmonics. Therefore, further attenuation D of more than 40 dB is needed see Figure 11 detail 1. With the equation we have already evaluated we can now calculate the needed value for the inductance L. To calculate the inductance L, we need an initial value for the capacitance C. Experiments have shown that 2 x 4.7 µF, i.e. twice the value of the input capacitance, is a sensible output value. This procedure has been verified experimentally and provides the most sensible output values. To have a more realistic value for the capacitor for the calculation we must take the DC bias effect into account (see REDEXPERT ). The capacitance therefore will be set to 70% (6.8 µF) for the two capacitors 885012209048 . D will be set to 55 dB to have here a good attenuation with enough margin (15 dB margin) to the limits: After implementing the designed input LC filter with Lf = 6.8 µH ( 744774068 ) and Cf ≈ 10 µF (2x 885012209048), nearly 60 dB of attenuation could be achieved. The selected inductor has no derating in the current range under consideration. Furthermore, no limits are exceeded at any frequency within the spectrum range of 150 kHz to 30 MHz. 7. SUMMARY & CONCLUSION In summary, using DC/DC power modules allows developers to profit from their high efficiency but does not remove the need for EMI filter design. Figure 13 shows a block diagram of the whole system, including the MagI³C filter concept, for overall EMC (radiated and conducted) compliance with the EMI IEC 55032 / CISPR32 standards, as well as to the surge and burst immunity standards IEC 61000-4-4 and IEC 61000-4-5. The emission reduction filter does not include protection against transient overvoltages, which must be added in practice. For this purpose, voltage-limiting components are added to the filter. A separate application note covers the details ( ANS023 ). For any application using a non-isolated DC/DC switching regulator, power module or not, EMI filtering must be considered. An improper EMI design leads to malfunctioning due to unwanted interferences of components in the application area. Also, other nearby electrical devices can be affected. Therefore, to reach EMI compliance it is mandatory to consider EMI filtering. This application note represents the best EMI counter measures, whether the non-isolated DC/DC converter is implemented discretely or as a compact package solution of a power module. Würth Elektronik offers comprehensive, competent technical support for all EMI filter designs as well as EMS protection designs including customer design-in support and layout reviews ( MagI³C Service & Support ). Figure 13: MagI³C filter concept for EMC compliance to IEC 55032 / CISPR32, IEC 61000-4-4 and IEC 61000-4-5. IMPORTANT NOTICE The Application Note is based on our knowledge and experience of typical requirements concerning these areas. It serves as general guidance and should not be construed as a commitment for the suitability for customer applications by Würth Elektronik eiSos GmbH & Co. KG. The information in the Application Note is subject to change without notice. This document and parts thereof must not be reproduced or copied without written permission, and contents thereof must not be imparted to a third party nor be used for any unauthorized purpose. Würth Elektronik eiSos GmbH & Co. KG and its subsidiaries and affiliates (WE) are not liable for application assistance of any kind. Customers may use WE’s assistance and product recommendations for their applications and design. The responsibility for the applicability and use of WE Products in a particular customer design is always solely within the authority of the customer. Due to this fact it is up to the customer to evaluate and investigate, where appropriate, and decide whether the device with the specific product characteristics described in the product specification is valid and suitable for the respective customer application or not. The technical specifications are stated in the current data sheet of the products. Therefore the customers shall use the data sheets and are cautioned to verify that data sheets are current. The current data sheets can be downloaded at www.we-online.com. Customers shall strictly observe any product-specific notes, cautions and warnings. WE reserves the right to make corrections, modifications, enhancements, improvements, and other changes to its products and services. WE DOES NOT WARRANT OR REPRESENT THAT ANY LICENSE, EITHER EXPRESS OR IMPLIED, IS GRANTED UNDER ANY PATENT RIGHT, COPYRIGHT, MASK WORK RIGHT, OR OTHER INTELLECTUAL PROPERTY RIGHT RELATING TO ANY COMBINATION, MACHINE, OR PROCESS IN WHICH WE PRODUCTS OR SERVICES ARE USED. INFORMATION PUBLISHED BY WE REGARDING THIRD-PARTY PRODUCTS OR SERVICES DOES NOT CONSTITUTE A LICENSE FROM WE TO USE SUCH PRODUCTS OR SERVICES OR A WARRANTY OR ENDORSEMENT THEREOF. WE products are not authorized for use in safety-critical applications, or where a failure of the product is reasonably expected to cause severe personal injury or death. Moreover, WE products are neither designed nor intended for use in areas such as military, aerospace, aviation, nuclear control, submarine, transportation (automotive control, train control, ship control), transportation signal, disaster prevention, medical, public information network etc. Customers shall inform WE about the intent of such usage before design-in stage. In certain customer applications requiring a very high level of safety and in which the malfunction or failure of an electronic component could endanger human life or health, customers must ensure that they have all necessary expertise in the safety and regulatory ramifications of their applications. Customers acknowledge and agree that they are solely responsible for all legal, regulatory and safety-related requirements concerning their products and any use of WE products in such safety-critical applications, notwithstanding any applicationsrelated information or support that may be provided by WE. CUSTOMERS SHALL INDEMNIFY WE AGAINST ANY DAMAGES ARISING OUT OF THE USE OF WE PRODUCTS IN SUCH SAFETYCRITICAL APPLICATION. DIRECT LINK ANS018 | EMI Filter Design for Non-Isolated DC/DC Converters USEFUL LINKS: Application Notes : https://we-online.com/en/support/knowledge/application-notes Services: https://we-online.com/en/products/components/service Contact : https://we-online.com/en/support/contact CONTACT INFORMATION Würth Elektronik eiSos GmbH & Co. KG Max-Eyth-Str. 1, 74638 Waldenburg, Germany Tel.: +49 (0) 7942 / 945 – 0 Email: appnotes@we-online.de Web: https://www.we-online.comdc/dc_converterwürth elektronikmagi³c_power_moduleapplication_noteknowledgewurth_elektronikhttps://community.element14.com/products/manufacturers/wuerth-elektronik/b/blog/posts/redcube-press--fit-terminals-for-automotive-applicationsWed, 30 Jul 2025 08:05:00 GMT93d5dcb4-84c2-446f-b2cb-99731719e767:8d6ecb16-e92e-406b-8d34-5d6078e7591eWürth ElektronikApplication Note REDCUBE PRESS- FIT Terminals for automotive applications SN025 by Markus Stark 1. THE PRESS-FIT TECHNOLOGY REDCUBE PRESS-FIT terminals stand out as the go-to choice for robust high-power connections at the PCB level in automotive applications. They're widely employed for connecting wiring with cable lugs to circuit boards. The current rating of REDCUBE PRESS-FIT is seriously impressive. Even with the same power output, these components generate the least amount of heat compared to alternatives for powering PCBs. Figure 1: PRESS-FIT Technology. By pressing the pins directly into the PCB, there's a strong friction between the pin and the plated through-hole, creating a seamless cold-welding connection between the pin and the copper-plated via within the PCB. This delivers a secure, gastight, and mechanically strong connection with a contact resistance of less than 200 µOhm. No other technology matches the ability to handle currents up to 250 A with such minimal self-heating. In terms of long-term reliability, REDCUBE PRESS-FIT surpasses a surface-mount technology (SMT) solder joint by up to 30 times. A single solid press pin typically requires a force of 100 N to extract it from a 1.6 mm PCB. So, even a small component with 8 pins could handle the weight of an average person without being pulled out of the PCB. This makes REDCUBE PRESS-FIT Terminals ideal for offering both electrical and mechanical connection solutions for electronic components. If after pressing the connector into the printed circuit board, the solid press pin interference on each of the four corners is greater than 3° against the via sleeve, the press connection zone exhibits lower electrical resistance than the brass pin itself. This ensures there's no electrical or thermal bottleneck. The example in Figure 2 shows a cross-section of the pressed pin in the via. Typically, the connection surface angle is significantly larger, offering ample safety margin for the electrical connection. Figure 2: Required connection angle only 3°. The 3° refers to the to a portion of a full circle’s 360° At the same time, this area should not exceed 12°, otherwise, the via copper plating can be damaged. The values specified in the data sheet ensure compliance with this requirement. Press-fit technology offers several advantages over solder technology. It's particularly adept at handling very thick circuit boards with heavy copper plating. Additionally, it allows for trouble-free two-sided mounting of circuit boards, facilitating compact module designs. With Press-fit technology, there's consistent contact between the pin and the copper layer along the entire length of the press-fit zone. This reliability is not assured with soldering, as solder may not fully rise along the entire length of the via, leading to increased transition resistances. Consequently, the long-term reliability and mechanical stability are not as robust as with REDCUBE PRESS-FIT. Unlike soldering, this process does not subject the circuit boards to thermal loading. REDCUBE PRESS-FIT connectors from Würth Elektronik are crafted from CuZn39Pb3 material, ensuring they comply with RoHS standards regarding copper alloys. 2. THE APPLICATION As already mentioned, REDCUBE PRESS-FIT Terminals are capable of carrying currents exceeding 250 A on the circuit board. However, it's crucial to evaluate the current-carrying capacity of REDCUBE PRESS-FIT within the context of the entire system. Factors such as conductor path thickness, width, cable cross-section, ambient temperature, and heat distribution must all be carefully considered when selecting the appropriate REDCUBE PRESS-FIT terminals for the application. The press-fit zone itself exhibits extremely low resistance, ranging from 100 to 200 µOhm. As a result, the limiting factor often arises from the layout of the connected conductor paths or the connection of external feed lines to a press-fitted component. Indeed, the challenge in designing high-current applications lies in achieving the optimal interaction among all components of the system. 3. THE PROCESSING Hot Air Leveling (HAL) as well as electroless nickel immersion gold (ENIG) PCBs are not recommended to use. HAL surfaces are inhomogeneous and reliable contacting across all layers cannot always be guaranteed. ENIG surfaces are very hard and brittle. This can lead to chip formation on the pin or to contact problems with the PCB in conjunction with the existing nickel layer. Immersion-coated PCB surface finishes, on the other hand, ensure even distribution of tin throughout the via hole. This makes it easier to meet tolerances and effectively prevents chip formation. Chips (plating or solder pushed out of the via) can affect the technical cleanliness of the system and, although rather unlikely, can lead to short circuits as a conductive particle. REDCUBE PRESS-FIT Terminals feature square-shaped press-fit pins. The design specifications for the through-hole are outlined in the respective specifications and must be adhered to. For optimal performance with a standard REDCUBE PRESSFIT, the circuit board thickness should ideally fall within the range of 1.6 to 2.7 mm. The typical extraction force is approximately 100 N per pin for a PCB thickness of 1.6 mm. Greater circuit board thicknesses correspond to correspondingly higher values. Figure 3: Specification for chemical surfaces. Mounting a copper rail to increase the current-carrying capacity is also achievable with REDCUBE PRESS-FIT terminals. This can be done in two ways: by press-fitting the copper rail beneath the circuit board or by screwing it onto the REDCUBE PRESS-FIT. During the press-fit process, it's important to ensure that the maximum overall thickness of the circuit board with the copper rail does not exceed 2.7 mm. Figure 4: Pressing process: PCB directly with copper bar. Furthermore, REDCUBE PRESS-FIT terminals are ideally suited for fulfilling purely mechanical functions, such as connecting circuit boards to cases or linking two circuit boards together. 4. THE DESIGN HINTS Other components should be mounted at least 4 mm from the press-fit hole due to slight PCB flexing during insertion. The press-fit hole should have a minimum distance of 3 mm from the edge. The press-fit force per pin should range from a minimum of 40 N to a maximum of 250 N. Typically, for a 1.6 mm PCB, this force falls around 100-150 N per pin. The exact values can be found in the respective datasheet. During the entire press-fit process, it's essential to support the press-fit zone. Without proper support, there's a risk of circuit board deflection during pressing. This is especially crucial for pneumatic presses, where it's important to ensure that the stroke cycle is performed evenly and consistently. 3D printed or aluminum supporters are only suitable for a few samples. The stroke cycle should be executed in one movement perpendicular and without angular misalignment to the PCB. This prevents defects in the via or delamination of the printed circuit board. Following press-fitting, the pins should slightly protrude from the circuit board and no components should press against them. The recommend minimum separation is 0.1 mm. Our REDCUBE PRESS-FIT terminals are specifically designed for press fitting. Alternative processing methods, such as soldering, are not recommended. Due to their high heat absorption, press-fitting of the REDCUBE PRESS-FIT should be carried out last, after all soldering processes are completed. Re-soldering REDCUBE PRESS-FIT terminals after the press-fit process is also not recommended. Re-soldering can potentially lead to partial destruction of the cold weld and delamination in the circuit board, resulting in the permanent loss of mechanical stability in the pressfit zone. To prevent mechanical damage to the REDCUBE PRESSFIT and the PCB, it's crucial to adhere to the maximum permissible torques. The exact values can be found in the respective datasheet. The force-fitting speed shall be a maximum of 250 mm/min. IMPORTANT NOTICE The Application Note is based on our knowledge and experience of typical requirements concerning these areas. It serves as general guidance and should not be construed as a commitment for the suitability for customer applications by Würth Elektronik eiSos GmbH & Co. KG. The information in the Application Note is subject to change without notice. This document and parts thereof must not be reproduced or copied without written permission, and contents thereof must not be imparted to a third party nor be used for any unauthorized purpose. Würth Elektronik eiSos GmbH & Co. KG and its subsidiaries and affiliates (WE) are not liable for application assistance of any kind. Customers may use WE’s assistance and product recommendations for their applications and design. The responsibility for the applicability and use of WE Products in a particular customer design is always solely within the authority of the customer. Due to this fact it is up to the customer to evaluate and investigate, where appropriate, and decide whether the device with the specific product characteristics described in the product specification is valid and suitable for the respective customer application or not. The technical specifications are stated in the current data sheet of the products. Therefore the customers shall use the data sheets and are cautioned to verify that data sheets are current. The current data sheets can be downloaded at www.we-online.com. Customers shall strictly observe any product-specific notes, cautions and warnings. WE reserves the right to make corrections, modifications, enhancements, improvements, and other changes to its products and services. WE DOES NOT WARRANT OR REPRESENT THAT ANY LICENSE, EITHER EXPRESS OR IMPLIED, IS GRANTED UNDER ANY PATENT RIGHT, COPYRIGHT, MASK WORK RIGHT, OR OTHER INTELLECTUAL PROPERTY RIGHT RELATING TO ANY COMBINATION, MACHINE, OR PROCESS IN WHICH WE PRODUCTS OR SERVICES ARE USED. INFORMATION PUBLISHED BY WE REGARDING THIRD-PARTY PRODUCTS OR SERVICES DOES NOT CONSTITUTE A LICENSE FROM WE TO USE SUCH PRODUCTS OR SERVICES OR A WARRANTY OR ENDORSEMENT THEREOF. WE products are not authorized for use in safety-critical applications, or where a failure of the product is reasonably expected to cause severe personal injury or death. Moreover, WE products are neither designed nor intended for use in areas such as military, aerospace, aviation, nuclear control, submarine, transportation (automotive control, train control, ship control), transportation signal, disaster prevention, medical, public information network etc. Customers shall inform WE about the intent of such usage before design-in stage. In certain customer applications requiring a very high level of safety and in which the malfunction or failure of an electronic component could endanger human life or health, customers must ensure that they have all necessary expertise in the safety and regulatory ramifications of their applications. Customers acknowledge and agree that they are solely responsible for all legal, regulatory and safety-related requirements concerning their products and any use of WE products in such safety-critical applications, notwithstanding any applications-related information or support that may be provided by WE. CUSTOMERS SHALL INDEMNIFY WE AGAINST ANY DAMAGES ARISING OUT OF THE USE OF WE PRODUCTS IN SUCH SAFETYCRITICAL APPLICATIONS. DIRECT LINK SN025 | REDCUBE PRESS- FIT Terminals for automotive applications USEFUL LINKS: Application Notes : https://we-online.com/en/support/knowledge/application-notes Services: https://we-online.com/en/products/components/service Contact : https://we-online.com/en/support/contact CONTACT INFORMATION Würth Elektronik eiSos GmbH & Co. KG Max-Eyth-Str. 1, 74638 Waldenburg, Germany Tel.: +49 (0) 7942 / 945 – 0 Email: appnotes@we-online.de Web: https://www.we-online.comredcubeswürth elektronikapplication_notewurth_elektronikhttps://community.element14.com/products/manufacturers/wuerth-elektronik/b/blog/posts/ans023-transient-protection-sip-3Mon, 14 Jul 2025 13:48:00 GMT93d5dcb4-84c2-446f-b2cb-99731719e767:7ad1883f-a814-40fd-849b-5d40e0ffa596Würth ElektronikApplication Note Transient protection SIP-3 ANS023 by Timur Uludag 1. INTRODUCTION AND THEORETICAL BACKGROUND Transient over voltages often occur in industrial environments due to an extensive electrical infrastructure. In order to develop an efficient filter to limit transient over voltages, many influencing parameters must be considered. Today, many industrial applications run on logic input voltage levels such as 5 VDC or lower, as shown in Figure 1. The power distribution system used to power such applications often run on a DC bus voltage of 24 VDC. Switching DC/DC converters are commonly chosen to handle the conversion of the higher DC bus voltage down to the lower logic voltage level. Figure 1 shows the basic electrical structure of an industry plant. The separated parts of the applications are supplied through a DC bus. On site, every separated electrical load is connected via a sub distribution with 24 V. Non-isolated power modules are implemented to provide the operating voltage of all subsystems. 1.1 Origin of Transients Figure 1: Industrial environments typically use a DC bus to power the individual elements of a plant Transient over voltages can be defined as short-term deviations from a nominal voltage value, which exceeds the permissible tolerance range of the nominal voltage in an electrical system. The effects of the transient over voltages are mostly destructive. First and foremost, there is more than one possible cause for a transient over voltage on the DC bus that leads to an abrupt rise. The origin of the over voltage can be a lightning strike (Figure 1, Part A), in which case we speak of a surge, or it can be generated in the system itself (Figure 1, Part B). Classically, the output of switched mode power supplies (SMPS ,DIN Rail power supply) specify surge voltage on output according to the EN 61000-4-5 between a positive and negative value ranging from 500 V to 1 kV. Since the output of the DIN rail power supply is connected directly to the input of the DC/DC converter, it needs to have the same test severity levels. Improper and nonexistent surge/transient protection leads to malfunctioning due to electrical damage to the DC/DC converter, leading to higher system downtimes and costs. For a correct calculation, we must use a standardized transient pulse as defined in the IEC 61000-4-5 standard as a so-called surge. 1.2 Filter Concept Figure 2: Filter concept for EMC compliance of MagI³C power modules with immunity and emission filter Figure 2 shows the entire immunity filter concept (green), which includes two stages of filtering. One stage to clamp the high transient overvoltage during a surge event, which can be achieved with transient suppressor components such as unidirectional TVS diodes. For the second stage, a passive LC filter is recommended, which attenuates the voltages that exceed the maximum operating voltage of the DC/DC converter. For optimization purpose the two filter stages could be merged into one filter stage that includes both, the immunity and the emission filter. This application note targets the step-by-step approach with two separate filter stages as this is the easiest. 2. IMMUNITY FILTER DESIGN LIMITS In many cases, two different values are specified in the data sheet in the chapters "ABSOLUTE MAXIMUM RATINGS" and "OPERATING CONDITIONS". The value in the "ABSOLUTE MAXIMUM RATINGS" one is the absolute maximum input voltage V PM_MAX , which, if exceeded, will result in permanent damage to the power module The other is the maximum operating voltage, which corresponds to the maximum input voltage specified by the manufacturer for which the power module is properly used. To protect against transient overvoltages, it is recommended that the immunity filter is designed so that the absolute input voltage of the power module never exceeds the absolute maximum input voltage V PM_MAX , even during transient overvoltage events. For further calculations the 173010535 SIP-3 power module with an absolute maximum input voltage of V PM_MAX = 44 V is used as an example [1]. 2.1 Immunity Filter Design In this article, the use of a unidirectional TVS-Diode is considered as a protection element for the power module input. Unidirectional operational behavior means that the V-I characteristics are nearly the same as that of a Zener diode. Consequently, the diode is normally used in reverse direction. Exceeding the component’s specific breakdown voltage causes the TVS diode to enter into a conductive state. The clamping voltage level is then determined by the current flowing through the component. The following numerical example provides a simplified, hands-on guide for establishing a filter. The filter estimation enables a quick refinement cycle when conducting real application tests. To make a proper design based on a TVS diode for a transient protection of the power module, the following parameters are required. V DC : supply voltage for the power module V BR : voltage where 1 mA of current is flowing through the TVS diode I PEAK : max. peak current flowing through the TVS diode @ V CLAMP_MAX P DISS : maximum dissipated power for the TVS diode V CLAMP_MAX : voltage where the diode carries the Ipeak 2.2 First Immunity Filter Stage Determination of V DC The maximum occurring DC bus voltage is decisive for the interference immunity of V DC , not the nominal value. With a 24 V bus, the voltage range of 19.2 to 30 V is specified in industrial environments. Consequently, the maximum value for V DC is 30 V. The selection of the TVS diode for the next steps of calculation is based on the available parts in the product portfolio of Würth Elektronik. The TVS diode 824550301 [2].is selected for the further calculation as it comes closest to the requirements. Determination of V BR V BR is defined as the voltage at which 1 mA of current is flowing through the TVS diode. This value, in this case 35.05 V, is not exactly fixed, as it is a PN junction whose voltage can vary slightly due to tolerances and operating conditions. The tolerance is given in the datasheet as ±5%. That leads to a V BR from 33.33 V to 36.80 V. In this region, the diode starts its conduction with a current of 1 mA. However, it is also necessary to know the value at which the transient voltage should be clamped. This is represented by the parameter V CLAMP_MAX . Determination of V CLAMP_MAX This value can be found in the datasheet as well. For the chosen diode, the voltage is 48.4 V based on a peak current of I PEAK of 31 A, representing a 10/1000 µs pulse. So far, the calculations have assumed an ideal laboratory environment with a controlled ambient temperature of 25°C. However, the reality looks different. From experience an ambient temperature around 55°C is common for electronic devices such as a TVS diode. Therefore, the calculation needs to be modified with a temperature factor. Mainly V CLAMP_MAX and the peak pulse power are strongly dependent on the temperature. Equation (1) shows the temperature effect to V CLAMP_MAX . (1) In the “standby case”, in which almost no current flows through the TVS diode apart from the leakage current of 1 µA, the junction temperature (Tj) is almost the same as the ambient temperature. Assuming a temperature coefficient αT for this type of TVS of 9.9 × 10-4 1/°C, this results in a maximum V CLAMP of 49.84 V at 55°C. This value will be now the starting point for the dimensioning of the second stage of the immunity filter. 2.3 Second Immunity Filter Stage The question now is how to estimate the right filter attenuation and how to get the best filter component values. Starting with the attenuation, the minimum filter attenuation can be calculated with Equation (2). (2) Instead of the symbol A (attenuation) the symbol G (gain) is used. A negative gain means an attenuation. Equation (2) considers the resulting clamping voltage level V CLAMP_MAX of the TVS diode during the surge event and the maximum operating voltage level V PM_MAX of the chosen converter. The task is to design the filter according to Figure 3, where a LC filter circuit needs to be added to the TVS diode. Figure 3: Equivalent circuit for second stage immunity filter calculation. The designer, allowing for the calculation of a corresponding filter capacitor, can select the inductor value. The reason for this is that the filter inductor is in series with the application, and therefore its resistance (RDC) causes unwanted losses. Therefore, the inductor with the smallest possible RDC value concerning the nominal maximum output current of the DC/DC converter should be selected. A WE-PD2 ( 744776112 ) with an inductance of 12 µH, an RDC of 336 mΩ and a rated current of 2.72 A was selected for this example filter design.The DC input resistance of the power module can be determined with the given input and output voltage, the output current as well as the efficiency during operation. Putting these parameters together, Equation (3) for the converter input resistance can be established. (3) Figure 3 shows the equivalent circuit where the TVS diode is represented as a simplified voltage source. The remaining portion of the circuit diagram for the EMC model consists of two LC filters for immunity (surge protection) and emission (EMI attenuation), the input capacitor of the DC/DC converter and the input resistance of the regulator. As this is an indoor application and therefore an indirect surge caused by the overvoltage occurs, the following assumptions and calculations are based on an 8/20 µs pulse as defined in the IEC 61000-4-5 standard. For further simplification, it is possible to omit C f_E and L f_E since this filter is designed to suppress disturbances at the switching frequency of the power module (Figure 4). Figure 4: Simplified equivalent circuit for second stage immunity filter calculation. The power module switches typically at 520 kHz. Looking at the frequency spectrum of the surge impulse, one can determine that the highest value of the noise voltage occurs at a frequency of f = 1 kHz. That’s why the over voltage is based on 1 kHz. To get the attenuation G of the filter, it is necessary to compare the output voltage to the input voltage of the system (Equation (4)). (4) With the previously calculated DC input resistance RPM in of the converter, Equation (5) can be used to establish the necessary capacitor value: (5) Here L f represents the immunity filter inductor, R PM the DC input resistance of the converter, C IN the input capacitor and C f_EMISSION the EMI filter capacitor of an input PI filter structure. The values can be found in the datasheet of the power module 173010535 . Assuming the use of a 12 µH inductor, the calculation result for the filter capacitor is C f = 218 µF. Checking the standard values for capacitors, a value of 220 µF ( 860010775018 ) is chosen as it is larger than the calculated value. A value less than the calculated capacitance would not provide sufficient attenuation of the filter. The final filter selected components are as follows: TVS diode = 824550301 , L f_I = 744776112 , C f_I = 860010775018. The influence of the temperature on the V CLAMP_MAX and thus on the value of the filter capacitor, is shown in Table 1. T able 1: Temperature influence overview The capacitor values have been calculated, but real capacitors are subject to relatively high tolerances. The actual capacitance value can vary by up to ±20 %. If the temperature dependence of the VCLAMP_MAX is not fully accounted for, a capacitor with too low a capacitance value may be selected. 2.4 Transient Protection and EMI In order to develop an efficient filter to limit transient over voltages, many influencing parameters must be considered. This is particularly important for an application in an industrial environment, since transient over voltages often occur here due to the extensive electrical infrastructure. The filter with transient protection enables efficient protection of the DC/DC converter module under consideration and simultaneously enables damping of high-frequency emissions. Figure 5 shows the complete EMC concept for the immunity and emission filter. The values for the components used, are shown in Table 2. Table 2: BOM for immunity and emission filter. Figure 5: EMC compliance filter concept of MagI³C power modules with immunity and emission filter. A APPENDIX A.1 Literature [1] Würth Elektronik data sheets for SIP-3 power modules: https://www.we-online.com/MAGIC-FDSM [2] Würth Elektronik data sheets for TVS diodes: https://www.we-online.com/WE-TVSP IMPORTANT NOTICE The Application Note is based on our knowledge and experience of typical requirements concerning these areas. It serves as general guidance and should not be construed as a commitment for the suitability for customer applications by Würth Elektronik eiSos GmbH & Co. KG. The information in the Application Note is subject to change without notice. This document and parts thereof must not be reproduced or copied without written permission, and contents thereof must not be imparted to a third party nor be used for any unauthorized purpose. Würth Elektronik eiSos GmbH & Co. KG and its subsidiaries and affiliates (WE) are not liable for application assistance of any kind. Customers may use WE’s assistance and product recommendations for their applications and design. The responsibility for the applicability and use of WE Products in a particular customer design is always solely within the authority of the customer. Due to this fact it is up to the customer to evaluate and investigate, where appropriate, and decide whether the device with the specific product characteristics described in the product specification is valid and suitable for the respective customer application or not. The technical specifications are stated in the current data sheet of the products. Therefore the customers shall use the data sheets and are cautioned to verify that data sheets are current. The current data sheets can be downloaded at www.we-online.com. Customers shall strictly observe any product-specific notes, cautions and warnings. WE reserves the right to make corrections, modifications, enhancements, improvements, and other changes to its products and services. WE DOES NOT WARRANT OR REPRESENT THAT ANY LICENSE, EITHER EXPRESS OR IMPLIED, IS GRANTED UNDER ANY PATENT RIGHT, COPYRIGHT, MASK WORK RIGHT, OR OTHER INTELLECTUAL PROPERTY RIGHT RELATING TO ANY COMBINATION, MACHINE, OR PROCESS IN WHICH WE PRODUCTS OR SERVICES ARE USED. INFORMATION PUBLISHED BY WE REGARDING THIRD-PARTY PRODUCTS OR SERVICES DOES NOT CONSTITUTE A LICENSE FROM WE TO USE SUCH PRODUCTS OR SERVICES OR A WARRANTY OR ENDORSEMENT THEREOF. WE products are not authorized for use in safety-critical applications, or where a failure of the product is reasonably expected to cause severe personal injury or death. Moreover, WE products are neither designed nor intended for use in areas such as military, aerospace, aviation, nuclear control, submarine, transportation (automotive control, train control, ship control), transportation signal, disaster prevention, medical, public information network etc. Customers shall inform WE about the intent of such usage before design-in stage. In certain customer applications requiring a very high level of safety and in which the malfunction or failure of an electronic component could endanger human life or health, customers must ensure that they have all necessary expertise in the safety and regulatory ramifications of their applications. Customers acknowledge and agree that they are solely responsible for all legal, regulatory and safety-related requirements concerning their products and any use of WE products in such safety-critical applications, notwithstanding any applicationsrelated information or support that may be provided by WE. CUSTOMERS SHALL INDEMNIFY WE AGAINST ANY DAMAGES ARISING OUT OF THE USE OF WE PRODUCTS IN SUCH SAFETYCRITICAL APPLICATION. DIRECT LINK ANS023 | Transient protection SIP-3 USEFUL LINKS: Application Notes : https://we-online.com/en/support/knowledge/application-notes Services: https://we-online.com/en/products/components/service Contact : https://we-online.com/en/support/contact CONTACT INFORMATION Würth Elektronik eiSos GmbH & Co. KG Max-Eyth-Str. 1, 74638 Waldenburg, Germany Tel.: +49 (0) 7942 / 945 – 0 Email: appnotes@we-online.de Web: https://www.we-online.comSIP-3würth elektronikapplication_noteknowledgewurth_elektronik